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Abstract
In this paper, an actively tunable rasorber with broadband RCS reduction and low infrared emissivity is proposed. The rasorber can achieve flexible control of the peak of the transmission frequency and make the platform invisible in multiple spectrum. Based on the combination of varactor diodes and bandpass frequency-selective surface (FSS), the transmission window can be continuously tuned from 1.8 to 4.5 GHz. The designed rasorber has more than 10 dB RCS reduction from 5.4 to 14.1 GHz. Furthermore, an infrared low emissivity layer based on ITO resistance film is added above the rasorber, and the average infrared emissivity of the measured surface is 0.33. The experimental and simulation results are in good agreement. This work is expected to be applied to frequency hopping secure communication and ultra-wideband, multi-spectrum stealth.
© 2023 Optica Publishing Group under the terms of the Optica Open Access Publishing Agreement
1. Introduction
Modern target detection is not limited to the radar band, but extended to the infrared and visible bands. Future platforms must be capable of multi-spectrum stealth [1–3]. However, the traditional absorber is limited to working in a single frequency band, which is difficult to meet the requirements of the application. Meanwhile, the frequency-selective rasorber (FSR) needs to transmit electromagnetic waves at a specific frequency, so it is necessary to design an efficient transmission window [4–6]. However, most of the current radar radome transmission windows are fixed and lack flexibility. Therefore, it is necessary to design an efficient transmission rasorber with active tuning [7–9], which can make the transmission frequency reconfigurable.
There are many reports on the traditional radar-infrared compatible stealth structure, which usually adds an infrared shielding layer on the surface of the lossy absorber. This infrared shield has a low infrared emissivity and a relatively high transmittance in the microwave band [1–3,10–14]. Radar waves can easily pass through the infrared shield without causing reflection and are absorbed in the absorber by generating ohmic loss. However, the heat generated by the ohmic loss of the absorber will increase the infrared radiation of the target to a certain extent, which will greatly increase the probability of the target being detected by the infrared detector [6]. In addition, the radar cross section (RCS) can also be reduced by coding metasurface based on the principle of phase cancellation [15,16]. By constructing unit cells with a phase difference of 180° and coding arrangement, the incident energy can be scattered in multiple directions, thus reducing the overall RCS. In addition, there are metasurfaces based on the integration of absorption and scattering [17–19]. By adding resistance loss or graphene resistance film, the combination of reducing the backward reflection amplitude and phase cancellation can further expand the stealth bandwidth and depth.
Radar radome should realize the function of electromagnetic wave transmission and stealth at the same time. In the transmit/receive frequency band of the antenna, the radome requires efficient transmission windows. While outside the antenna operating band, low reflectivity is often required to prevent detection by radar. There are many kinds of rasorber at present, such as the combination of high-frequency absorption and low-frequency transmission [8,20], that is, the transmission window and the absorption band are not at the same frequency. Besides, there are transmission windows in the absorption frequency band, such rasorber can often achieve broadband stealth and transmit at specific frequencies [21–25]. In addition, the actively tunable rasorber is also a hot research topic, and the working state of such rasorber is not fixed, but can be flexibly tuned. For example, by adding a PIN switching diode, flexible switching of reflection and transmission modes can be achieved by conducting and cutting the switch [4,8,9,26]. In addition, there are reports of varactor-diode-based rasorber, which can realize the continuously tunable frequency of the transmission peak within a certain range, and then realize the transmission and communication in multiple frequency bands [7,27–30].
In order to achieve flexible control of the electromagnetic wave transmission frequency band and make the platform invisible in multiple spectrums, this paper proposes an actively tunable rasorber with wideband RCS reduction and low infrared emissivity. Based on the combination of varactor diodes and bandpass frequency-selective surface (FSS), the transmission window can be continuously tuned from 1.8 to 4.5 GHz. All cells are fed simultaneously through metal vias and underlying feeders. Furthermore, in order to realize the radar stealth function of the rasorber, the I-shaped pattern is adopted as the polarization conversion unit. By rotating the I-pattern 90 degrees, two 1-bit coding units with 180° phase difference are formed. In addition, we adopt Simulated annealing algorithm (SAA) to optimize the far field pattern of the rasorber for plane wave incident, and then design the coding arrangement with the lowest RCS to realize the wideband stealth of rasorber. The designed rasorber has more than 10 dB RCS reduction from 5.4 to 14.1 GHz, and perfect angular stability within 30° of the incident angle. Next, we added an infrared low emissivity layer based on ITO resistance film on the rasorber surface, and the infrared emissivity of the test surface is 0.33. Due to the phase cancellation property of the scattering surface for the incident electromagnetic wave, almost no ohmic heat is generated, which also guarantees the low infrared emissivity of the rasorber and achieves multifunctional compatibility. Finally, we fabricated a 300mm × 300 mm sample for experimental measurement, and the experimental results are in good agreement with the simulation.
2. Design and analysis of the actively tunable rasorber
2.1 Design process of overall structure
The proposed rasorber with wideband RCS reduction and low infrared emissivity is shown in Fig. 1. The whole structure consists of three layers with a unit period of 15 mm. The bottom layer is a tunable transmission-type rasorber, as shown in Fig. 2(a). The bottom layer is an F4B dielectric substrate with a dielectric constant of 2.2 and a thickness of t1 = 2.5 mm. The upper surface of the substrate is a metal pattern, the periphery is a square ring structure, and the middle square patch is rotated by 45°. The square patch and the outer ring are connected by four varactor diodes, and the width of the slit is 0.25 mm. The model of the varactor diode is MAVR-000120-1411. When the reverse bias voltage changes from 0-15 V, the capacitance of the varactor-diode changes from 0.2-1.2 pF [31], so as to realize the frequency switching of the transmission peak. The average internal resistance of the varactor-diode is 0.88Ω, and the ohmic loss introduced is as small as possible in order to minimize the insertion loss and improve the transmission. The lower surface of the F4B substrate is distributed with a cross-shaped feeder line and a periodic square ring structure. The feeder is slit in the middle and 12 inductors are added to insulate unnecessary induced current. The inductor model is LQW15AN75NG80D, the inductance value is 75nH, and the self-resonant frequency is 3.62 GHz. The series of multiple inductors makes the feeder achieve high impedance characteristics in the frequency-modulated range, which is conducive to the realization of the transmission function. The thin square ring of the bottom layer is a number of band-stop units, which can achieve good out-of-band suppression characteristics.
Fig. 1. Schematic diagram of the overall unit structure. (Parameters: L = 15 mm, h1 = 4 mm, h2 = 0 mm, t1 = 2.5 mm, t2 = 0.5 mm, t3 = 0.125 mm.)
Fig. 2. Detail diagram of each layer. (a) Tunable transmission layer. (b) Infrared stealth layer. (c) Scattering layer. (Parameters: P1 = 0.85 mm, P0 = 0.15 mm, S1 = 6 mm, S2 = 7.7 mm, S3 = 2.3 mm, S4 = 2.3 mm, S5 = 1.5 mm, S6 = 2.5 mm, S7 = 3 mm, L1 = 5.65 mm, L2 = 11.8 mm, L3 = 6 mm, L4 = 0.7 mm, W1 = 3.95 mm, W2 = 1.5 mm, W3 = 0.5 mm, W4 = 1.4 mm, W5 = 0.3 mm.)
For efficient feeding, the upper metal patch is connected to the bottom feeder through the metalizing vias, which pass through the intermediate dielectric layer. The metal patch and the bottom feeder act as positive and negative electrodes, respectively, and are fed together through an external voltage source. The varactor diodes in all cells can obtain the same bias voltage, thus achieving the tunable property of rasorber. This design can simplify the feeding network and minimize the adverse effect of the feeding network on the rasorber.
In addition, we design a scattering unit in the middle layer to achieve the broadband stealth of rasorber, as shown in Fig. 2(c). The middle layer is a figure-I scattering unit with a medium of F4B and a thickness of 0.5 mm. Between the middle layer and the bottom layer is air with a thickness of h1 = 4 mm. The medium surface is covered with an I-shaped metal pattern with four slight bumps in the middle of the pattern. The addition of scattering elements has almost no effect on the underlying transmittance. The I-shaped pattern is used as the polarization conversion unit. By rotating the I-pattern 90 degrees, two 1-bit coding units with 180° phase difference can be formed. Furthermore, the Simulated annealing algorithm was used to optimize the far field pattern of the rasorber for plane wave incident, and then the coding arrangement with the lowest RCS was designed to achieve the wideband stealth performance of the rasorber. Then, we added the infrared stealth layer on the top layer, as shown in Fig. 2(b). The substrate is a 0.125 mm thick transparent PET medium covered with periodic ITO square patches. The square resistance of the ITO resistance film is 6 Ω/sq, and the gap between the ITO patches is 0.15 mm. The overall thickness of rasorber is 7.125 mm. This design can ensure very low infrared emissivity and very high microwave transmittance to reduce the adverse effects on the function of the underlying structure [14].
2.2 Equivalent circuit model (ECM) analysis
The equivalent circuit of the proposed rasorber is shown in Fig. 3. Firstly, the outermost infrared stealth layer consists of periodic ITO square patches, and there is mainly a capacitance effect between each patch. Therefore, the infrared stealth layer behaves as a low-pass filter in the microwave band [12], so it is equivalent to the capacitance C1. Secondly, the I-shaped patch of the scattering layer can be equivalent to a series LC structure, denoted by L2 and C2, respectively. For the tunable transmission layer, the outer metal square ring is equivalent to the parallel inductance L3, and the equivalent parallel coupling capacitance C3 exists between the central square patch and the square ring. The varactor, as a tunable element, is connected in parallel to the intrinsic circuit of the structure. Therefore, the varactor diode can be equivalent to a parallel variable capacitor Cv. In addition, the varactor diode also has parasitic resistance [33], so it is equivalent to a series resistance with R = 0.8Ω. Therefore, the tunable transmission layer is a parallel LC structure, which exhibits the characteristics of bandpass filtering. Moreover, the middle square patch has an extra inductance effect, and this inductance is in series with the bandpass structure, so it can be equivalent to the inductance L0. The periodic thin square ring at the bottom is the band stop unit, which can achieve good out-of-band suppression characteristics. Then the underlying square ring is equivalent to the series structure of L4 and C4. The medium between the layers can be equivalent to a transmission line of equal length, and the characteristic impedance of the medium can be defined as $Z = {\eta _0}/\sqrt {{\varepsilon _{r}}} $, where ${\varepsilon _{r}}$ is the dielectric constant of the medium and ${\eta _0}$ is the free space wave impedance. Therefore, the characteristic impedance ZP of PET is 217Ω, the characteristic impedance ZF of F4B is 254Ω, and the characteristic impedance ZA of air is 377Ω. The equivalent admittance of each layer structure is defined as follows:
Fig. 3. Equivalent circuit model (Parameters: C1 = 0.02pF, C2 = 3.7pF, C3 = 0.02pF, Cv = 0.2-1.2pF, C4 = 1.15pF, L0 = 2.1nH L2 = 11.5nH, L3 = 11.2nH, L4 = 44nH, R = 0.8Ω, h1 = 0.125 mm, h2 = 0.5 mm, h3 = 4 mm, h4 = 2.5 mm.)
According to the transmission line model theory, the whole rasorber can be equivalent to a two-port network, as shown in Fig. 3. The corresponding S-parameters and the ABCD matrix for a two-port network are shown below:
According to the transmission line theory, in the case of open circuit, the transmission layer can obtain an ideal passband with low insertion loss, which means that the impedance Z3 at fT should be infinite, that is ${Z_3} \to \infty $, ${Y_3} \to 0$. In this case, we calculated that the influence of inductance L0 on the resonant frequency is small and almost negligible, so the resonant frequency can be approximately calculated as follows:
At the frequency of fT, the rasorber exhibits a high transmittance mode, while out of band, it exhibits a reflection mode. Since the distance between the central square patch and the peripheral metal square ring is large, its equivalent parallel coupling capacitance C3 is relatively small. This makes the varactor diode Cv the dominant factor in the total capacitance variation, which is beneficial to achieve a larger tunable range. Moreover, the introduction of the additional series inductance L0 corresponds to the valley of the transmission coefficient. This makes the ECM out-of-band suppression effect basically consistent with the full-wave simulation, and the value of the additional inductance L0 fitted by the ECM is 2.1nH.
The commercial electromagnetic simulation software CST Microwave Studio is used to simulate the designed rasorber, and periodic boundary conditions are set to simulate infinite planar structures. At the same time, the equivalent circuit in Fig. 3 is built in the circuit simulation software Advanced Design System (ADS), and the relevant circuit parameters are optimized and fitted. The results of the full-wave simulation and circuit calculation are shown in Fig. 4. As shown in Fig. 4(a), by changing the capacitance of the varactor diode, the transmission peak can be continuously tuned from 1.9 to 4.1 GHz. CST full-wave simulation results show that the insertion loss is -1.6 dB at 1.9 GHz and -1.4 dB at 4.1 GHz. The ADS circuit simulation results show that the insertion loss is -1.5 dB at 1.9 GHz and -0.5 dB at 4.1 GHz. The transmission coefficient of full-wave simulation is slightly lower than that of circuit simulation at 4.1 GHz. The possible reason is that the coupling between the unit cells is considered in the full-wave simulation, so the insertion loss becomes larger. The introduction of an additional series inductance L0 corresponds to the valley of the transmission coefficient. Figure 4(b) shows the simulation results of the reflection coefficients. When the capacitance value changes, the peak frequency of the reflection coefficient in the two cases basically coincides. However, the reflection coefficient of the full-wave simulation is lower, and this difference is within an acceptable range. The full-wave simulation results are in perfect agreement with the circuit calculation results, which also verifies the correctness of the equivalent circuit model.
Fig. 4. S-Parameters simulation results of tunable element structure. (a) Transmission coefficient S21. (b) Reflection coefficient S11.
3. Design and optimization of scattering structure with low infrared emissivity
Based on the previously designed cell structure, the I-shaped patch of the scattering layer is rotated by 90° to form two 0-1 coding cells with polarization conversion function, as shown in Fig. 5. In Fig. 5(a), when an x-polarized electromagnetic wave Ei is incident, it can be decomposed into two orthogonal components in u direction and v direction, namely Eiv and Eiu. According to the full-wave simulation results, the amplitude and phase of the reflected wave in the u direction are unchanged. However, the reflected wave in the v direction has a phase difference of 180° compared to the incident wave, so the direction is opposite. The reflected waves in the u and v directions are synthesized as Er, where Er is a y-polarized electromagnetic wave. So the unit has the function of polarization conversion. Similarly, in Fig. 5(b), a y-polarized reflected electromagnetic wave is also synthesized. Moreover, the amplitude of the reflected electromagnetic waves of the two 0-1 cells is equal, and the phase difference is 180°.
Fig. 5. Principle of polarization conversion. (a) Unit cell ‘0’. (b) Unit cell ‘1’. (c) Simulated S-parameter characteristics of polarization conversion unit cell.
The S-parameters of the rasorber for full-wave simulation is shown in Fig. 5(c). The capacitance value of the varactor diode is set to 0.8pF. Since the magnitude characteristics of 0-1 cells are completely consistent, only the magnitude characteristics of one cell are shown. It can be seen from Fig. 5(c) that this rasorber has an efficient transmission window at 2.3 GHz with an insertion loss of -0.6 dB. Meanwhile, from 4.9 to 14.2 GHz, the rasorber can achieve cross-polarized reflection greater than -2 dB, and co-polarized reflection coefficients are all less than -10 dB. In addition, the phase difference of the cross-polarized reflected waves of the two cells can be kept at about 180° all the time because there is a geometric relationship of 90 degrees rotation between 0-1 cells. While the phase difference at the transmission peak is not 180° because the majority of the power at this time is transmitted rather than reflected.
Since the 0-1 coding unit has efficient reflection characteristics from 4.9 to 14.2 GHz, the reflection amplitude is all approximated to one. Such a pattern of far-field scattering depends mainly on the phase response of the element. In order for the phase gradient between the elements to control the wavefront effectively, the size of the elements must be comparable to the wavelength. Therefore, the 0 and 1 cells are arranged by 2 × 2 to form supercells, and the size of each supercell is 30 mm. The far-field array factor of the encoding metasurface can be calculated as follows [32]:
In order to achieve wideband RCS reduction, the Simulated annealing algorithm (SAA) is used to calculate the optimal coding arrangement. The flow chart of the algorithm is shown in Fig. 6. The objective of the optimization is $Goal = \min [{\max (AF(\theta ,\varphi ))} ]$. In this way, the peak value of the far-field scattering function can be minimized and the scattering energy is more evenly distributed to reduce the backward RCS. In order to take the bandwidth into account, six frequency points between 6 GHz and 18 GHz are calculated and averaged as the objective function for optimization. The initial temperature is set to 1000°C, the cooling coefficient α=0.95, and the stable temperature is 1°C. The new solution is obtained by mutating each cell with a certain probability P0 = 0.2.
The acceptance probability P of Metropolis criterion is introduced to avoid the iteration falling into a locally optimal solution. The probability of accepting a bad solution decreases as the number of iteration steps increases (the temperature decreases), and finally the optimal solution is found. The final obtained encoding arrangement is shown in Fig. 7. Figure 7(c) shows the coding matrix optimized by MATLAB, where blue and yellow represent 0 and 1 cells, respectively. Figure 7(a) shows the unit arrangement of the whole board, and Fig. 7(b) shows the detailed magnification of the supercell.
Fig. 7. (a) Unit arrangement of the whole board. (b) Detail magnification of the supercell. (c) Coding matrix optimized by MATLAB.
The far-field RCS of the rasorber whole plate is simulated in the commercial electromagnetic software CST Microwave Studio, and the RCS reduction value is calculated by subtracting it with the RCS of the same size perfect electrical conductor (PEC). As shown in Fig. 8, there is a greater than 10 dB RCS reduction from 8 to 13 GHz, allowing full coverage of the X-band. And there is greater than 8 dB RCS reduction from 7.6 to 14.2 GHz. Moreover, in the low-frequency band, the RCS reduction also shows a trough value corresponding to the low frequency transmission window. As the varactor diode capacitance varies from 0.2 pF to 1.2 pF, the reflection coefficient of high frequency is almost unchanged. However, the reflection coefficient of the low frequency will have a certain degree of jitter, which is caused by the transmission peak generated at the low frequency.
4. Experimental measurement and results
In order to verify the effectiveness of the simulation results of the tunable rasorber, a 300mm × 300 mm sample was fabricated using standard PCB technology for testing. As shown in Fig. 9(a), both the transmission and scattering layers were fabricated using F4BM220 substrate (εr = 2.2, tan σ = 0.0015). For the underlying tunable transmission layer, there are a total of 20 × 20 identical cells. For each unit, four varactor diodes are soldered using surface mount technology, and the varactor model is MAVR-000120-1411. For the scattering layer, there are a total of 10 × 10 supercells. PMI foam with a low dielectric constant was used to replace the air layer, the dielectric constant was 1.03, and the thickness of PMI was 4 mm. The top layer is a transparent ITO resistance film, which acts as an infrared stealth layer. The feeding structure uses the upper metal patch as a positive electrode and induces circular vias. The metal feeder on the back is used as the negative electrode, and the inductor is welded in the middle, and the inductor model is LQW15AN75NG80D. The external voltage source is connected to the positive and negative electrodes separately, so the varactor diodes in all cells can obtain the same bias voltage, thus achieving the tunable characteristics of the rasorber. As shown in Fig. 9(b). The ITO film is pasted on the surface of the scattering layer, and the overall structure is fixed by nylon clips, as shown in Fig. 9(c).
Fig. 9. (a) Layered presentation of rasorber. (b) Backside feeder diagram. (c) The overall structure after assembly.
The transmission and reflection characteristics of the rasorber were measured separately using the free space method. As shown in Fig. 10(a), the transmission performance of the rasorber is measured through the absorber screen. A pair of broadband horn antennas are placed on the front and back of the absorbing screen to transmit and receive electromagnetic waves. Rotating the screen allows for measuring the transmission performance of oblique incidence. The transmission coefficient S21 was measured using a vector network analyzer (Agilent N5244B) connected to the horn antenna. As shown in Fig. 10(b), the reflection performance of the rasorber is measured in the arch frame. The bistatic RCS performance of the rasorber can be measured by varying the angle of the transmit/receive antennas. We adopt the bistatic RCS with an incidence angle of 5° to approximate the case of normal incidence. The two stages of the voltage source are connected to the positive and negative poles of the rasorber to provide a tunable bias voltage in the range of 0-15 V.
Fig. 10. (a) Experimental scenario of measuring transmittance through the absorbing screen. (b) The experimental scene of bistatic RCS measured by the arched frame.
Figure 11 shows the measured results of transmission coefficient. Under normal incidence, the transmission peak is continuously tunable from 1.8-4.5 GHz, the lowest transmission coefficient is -3.9 dB, and the highest transmission coefficient is -1.2 dB, as shown in Fig. 11(a). When the incidence angle is less than 30°, the transmission coefficient is almost constant, so it has good angular stability. At an incidence angle of 45°, the insertion loss becomes significantly larger. This is attributed to the strong scattering from the metal vias, since the electromagnetic wave at oblique incidence has a large electric field component along the direction of the metal vias, resulting in reduced transmission efficiency. As shown in Fig. 11(b), the transmission performance of TE and TM polarization is basically similar. Compared with the simulation results, the tunability performance is almost the same. However, the measured insertion loss is significantly larger, which is on average about 2 dB larger than the simulation results. Especially when the reverse bias voltage is low, a higher deviation will occur. This is due to an increase in the series parasitic resistance of the varactor diodes at lower bias voltages, which results in some deterioration of the transmittance [33]. In addition, the higher the bias voltage, the smoother the transmission peak while the transmission peak is steeper when the bias voltage is lower. This is due to the nonlinear relationship between the varactor diode and bias voltage [31].
Fig. 11. (a) The transmission coefficient of the measured TE mode. (b) The transmission coefficient of the measured TM mode.
The reflection coefficient of the designed rasorber in the normal incidence case is shown in Fig. 12(a). It can be seen that with the reverse bias voltage varying from 2-12 V, the reflection coefficient is less than -10 dB at 5.4-14.1 GHz, which can completely cover the X-band. And the reflection coefficient is less than -7 dB at 5.2-18 GHz, so it also has a good stealth effect in higher frequency bands, thus achieving broadband radar stealth. Moreover, the reflection coefficient also shows several valleys in the low frequency band, which correspond to the tunable transmission peak. The curve trend of the experimental results is almost the same as the simulation results, and the reflection coefficient is slightly lower than that of the simulation results. The reason for the improvement of the measured RCS reduction bandwidth compared to the simulation should be that the actual 0-1 cell has a certain loss and the reflection amplitude is less than one. As a result, the backward reflection energy is reduced, the RCS reduction bandwidth becomes wider and the RCS reduction depth becomes deeper. Figure 12(b) shows the reflection coefficient for oblique incidence at a bias voltage of 12 V. It can be seen that the reflection coefficient gradually increases as the incidence angle increases. The possible reason is that in the case of oblique incidence, the 0-1 element no longer satisfies the 180° phase difference, resulting in the inability to achieve effective diffuse scattering. However, when the incidence is 30°, the reflection coefficient is less than -6 dB in the target frequency range, and it still has a good RCS reduction effect. Therefore, the structure still has good angular stability. The overall thickness of the rasorber is only 7.125 mm, which is equal to 0.12 λM, where λM is the maximum operating wavelength. The experimental results are in good agreement with the simulation design, which also verifies the effectiveness and practicability of the proposed rasorber.
Fig. 12. (a) The measured reflection coefficient under normal incidence. (b) The measured reflection coefficient under oblique incidence (Bias 12 V).
Finally, the Fourier transform infrared spectrometer (iS50 FTIR) is used to measure the infrared radiation characteristics of the sample, as shown in Fig. 13. When the surface of the sample is not covered with the infrared stealth layer, the average infrared emissivity is 0.92 at 2.5-15 µm. However, when the surface of the sample is covered with the ITO-based infrared shielding layer, the average infrared emissivity is reduced to 0.33 at 2.5-15 µm. Therefore, the designed rasorber has excellent infrared stealth performance. In addition, since the stealth in the microwave band is based on the phase scattering principle, almost no additional ohmic loss heat is generated, which also guarantees a low infrared emissivity at operation.
Fig. 13. (a) Experimental scene of Fourier infrared spectrometer measuring infrared emissivity. (b) Infrared emissivity measurement results.
Table 1 presents the comparison between this work and previous work. The advantage of this work mainly lies in the wider continuously tunable bandwidth, and the insertion loss is almost the same as the previous work. In addition, the stealth bandwidth of the proposed rasorber completely covers the X-band, but slightly narrower than that of the ultra-wideband circuit analog absorber. It also meets the requirements of lightness and thinness. Furthermore, this work integrates the stealth of multiple spectrums as well as transmission and reflection together, which can achieve multi-function compatibility.
5. Conclusion
In this work, an actively tunable rasorber with broadband RCS reduction and low infrared emissivity is proposed. The transmission frequency can be continuously tunable from 1.8 to 4.5 GHz by applying a bias voltage of 0-15 V. In addition, the arrangement of coding metasurface calculated by the intelligent optimization algorithm can achieve wideband stealth at 5.4-14.1 GHz. And the reflection coefficient is less than -7 dB at 5.2-18 GHz, thus achieving broadband RCS reduction efficiently. Moreover, the measured infrared emissivity is only 0.33. Due to the phase cancellation property of the scattering surface, almost no ohmic heat is generated, which also guarantees the low infrared emissivity of the rasorber and achieves multifunctional compatibility. This work is expected to find applications in secure communication as well as in ultra-wideband and multi-spectrum stealth.
Funding
Fundamental Research Funds for the Central Universities (021014380186); National Natural Science Foundation of China (62071215, 62271243); Priority Academic Program Development of Jiangsu Higher Education Institutions; Jiangsu Provincial Key Laboratory of Advanced Manipulating Technique of Electromagnetic Wave.
Disclosures
The authors declare no conflicts of interest.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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