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A novel high-speed Mach-Zehnder modulator (MZM) fully integrated into a 90 nm CMOS process is presented. The MZM features ‘double-pass’ optical phase shifter segments, and the first use of integrated inductors in a ‘velocity-matched’ distributed-electrode configuration.
© 2015 Optical Society of America
1. Introduction
Monolithic integration of photonic and electronic circuits in a silicon die holds promise for dramatic cost reduction in optical transceivers [1]. CMOS Integrated Nano Photonics (CINPs) could bring traditional semiconductor industry efficiency and low cost manufacturing to optical interconnect applications. IBM’s sub-100nm CINP technology (CMOS9WG) is designed for multi-channel short reach optical interconnects up to 25 Gbaud rates, where optical components are monolithically integrated into the CMOS circuit analog and mixed-signal front end. This minimizes processing steps, mask levels, component parasitics, and packaging/assembly complexity [2]. However, technology solutions range from the hybrid approach [3–10], where electrical and optical components are on separate chips, to a fully monolithically integrated solution [1,2,11–16]. Hybrid approaches allow individual optimization of electronic and photonic elements. Monolithic platforms naturally facilitate co-optimization via a single electronic/photonic design environment [2] and enable wafer/chip test and disposition prior to assembly to optimize yield, and minimize cost. Furthermore, if CMOS9WG technology is used in hybrid applications it provides powerful functionality to optimize hybrid performance such as on chip decoupling capacitors, inductors, and e-fuse enabled component tuning to improve performance and yield after assembly, which ultimately could open this technology to a broader range of applications.
Significant progress has been reported on the design and manufacturing of CMOS compatible Mach-Zehnder modulators (MZMs) and free carrier plasma dispersion-based electro-optic phase shifters [16–42]. Silicon photonic MZMs give relatively good temperature stability, but when reverse bias depletion-based PN diode optical phase shifters are used devices generally become long, requiring traveling wave (TW) electrode designs and a relatively large footprint [16, 21, 29–31,42]. The TW MZM is designed such that the electrical radio frequency (RF) driver connected to the MZM electrode sees a distributed line capacitance from the long modulator electrode, instead of a large lumped capacitance. The line capacitance of the MZM electrode is directly related to the electrode characteristic impedance the RF driver sees as shown in Eq. (1). Proper component design minimizes the RF impedance mismatch and minimizes RF reflections, which typically improves performance. The TW MZM configuration also attempts to reasonably match the velocity of a modulating RF signal traveling along its electrode length with that of the light wave being modulated within the device. The microwave velocity and characteristic impedance are governed by Eqs. (1) and (2).
where ZRF is the RF characteristic impedance, vRF is the RF group velocity, L is the line inductance, and CRF is the frequency dependent RF line capacitance of the electrode. In addition, MZM electrodes can be periodically loaded with various electrode elements to create an effective electrode line capacitance, inductance, and impedance as long as the structures used in the periodically loaded electrode are significantly smaller than the RF wavelength within the device. For example, if a reverse bias plasma dispersion PN diode is used in a TW MZM, often the line capacitance of the diode junction is too high to achieve a characteristic impedance near 50 Ω. Proper velocity matching within the MZM can also be difficult to achieve. In this case lumped element inductors can be inserted periodically into the electrode to increase the effective inductance, as long as both the inductors and the length of the electrode segments between inductors are small relative to the RF wavelength of interest. However, the inductors also impact the RF group velocity and add frequency dependent RF loss, and so proper design requires a careful consideration of impedance and RF propagation and loss characteristics within the periodically loaded MZM. We call such an MZM, with periodically inserted lumped RF elements, a distributed electrode MZM design.The ability to monolithically integrate electro-optic (EO) modulators into a CMOS manufacturing platform with multiple back-end metal layers fundamentally changes the design space available for optimizing transmitter performance. In this report, we present a novel distributed electrode MZM with periodically-placed monolithically-integrated radio frequency (RF) inductors used in conjunction with double-pass optical phase shifter segments. The double-pass optical phase shifter segments have the potential to cut the required MZM footprint in half, while the use of integrated inductors facilitates excellent impedance matching to 50 Ω and significantly improves RF/optic velocity matching within the MZM.
2. Mach-Zehnder modulator design
The MZM was designed with a distributed lumped-element concept, as shown in Fig. 1(a), where the electrode has concatenated capacitive and inductive lumped-elements to achieve the desired transmission characteristic impedance and RF velocity. The capacitive elements are depletion-based PN diode optical phase shifters. Figure 1(b) shows schematically how the optical waveguides are configured for ‘double pass’ interaction with the RF signal in the MZM. An image of our distributed electrode MZM, which was fabricated in IBM’s CMOS9WG technology, is shown in Fig. 1(c). Figure 1(d) shows an expanded view image with schematic highlights to help convey the design methodology. The distributed electrode MZM has a series of 100-μm-long RF/optic interaction sections that contain two 100 μm Si rib waveguide PN junction optical phase shifters, each being wired from signal to ground on both sides of the central signal electrode. The white arrow in Fig. 1(d) shows how the RF signal passes through each PN junction loaded electrode segment only once. However, the red arrows in Fig. 1(d) illustrate that the optical path does a ‘double-pass’ through each PN junction loaded segment. The light and RF fields co-propagate for one pass through the PN junction, and counter-propagate for the other. For high bandwidth each PN junction segment must be short enough to have a lumped element response, since the RF and optic fields counter propagate at certain times in the device. One 385 pH inductor is periodically integrated into the electrode for every 400 μm of optical PN junction length, which corresponds to 200 μm of loaded electrode length. Although the MZM footprint has not necessarily been minimized in this demonstration, the optical double-pass design efficiently utilizes the RF electrode length, and can significantly shrink the MZM footprint.
Fig. 1 a.) Schematic of the distributed electrode design segment with concatenated capacitive and inductive lumped-elements. b.) Schematic showing how the optical waveguides are configured to realize a ‘double-pass’ RF/optical interaction within the electro-optic phase shifters. c) Distributed electrode MZM with 0.8 mm of RF/optical interaction length on each arm of the push-pull MZM (1.6 mm total), d) Magnified view of distributed electrode ‘double pass’ design with schematic highlights.
The integrated inductors serve several purposes. First, they facilitate a 50 Ω impedance match in the distributed electrode. Each PN junction has a 0.65 pF/mm line capacitance when a −1V bias is applied. Note, however, that both sides of the signal line are loaded, so the actual line capacitance of the loaded electrode segment is 1.3 pF/mm. This large capacitance requires additional inductance to realize a 50 Ω effective electrode characteristic impedance. Second, since light makes a double pass through each RF/optic interaction section, while the RF signal makes a single pass, the light takes longer to travel through the loaded MZM segment than the RF signal. Inductors give extra RF delay so RF and optical signals traverse the MZM in a more closely matched manner. Furthermore, incorporating inductors into the MZM allows the use of optical phase shifters with extremely low Vπ·L products, which are typically associated with high PN junction line capacitance. The inductors enable the MZM to retain good RF propagation and impedance characteristics with high capacitance PN junctions, and using a lower Vπ·L product phase shifter further contributes to reducing MZM footprint by reducing the RF/optic interaction length required in the MZM.
The PN diode phase shifters have lateral junctions placed in the center of the waveguides and nominal peak N and P doping of 1.5 × 1018 cm−3. The SOI is 220 nm thick with 0.5 μm wide waveguides for operation with 1.5 μm light. The measured PN junction Vπ·L is 0.9 V-cm, for a 0 to −2V bias change and is ~1.4 V-cm for a 0 to −5V bias change, while the optical propagation loss is ~33 dB/cm at 0 V bias and ~31 dB/cm at −5 V.
3. Device response
Measurements were taken with high speed RF probes at the input and output of the RF electrodes, and the output probe was terminated with 50 Ω for bandwidth measurements. Two types of devices were investigated. The first had inductors incorporated into the MZM electrode design, as shown in Fig. 1(c), while the second was nearly identical but had no integrated inductors, as shown in Fig. 2(a). An MZM optical insertion loss of ~4 dB was measured for both the devices shown in Figs. 1(c) and 2(a). Since the PN junction optical loss was measured to be 33 dB/cm, and there are 0.8 mm of the phase shifters in each MZM arm, this indicates that ~2.7 dB of the MZM insertion loss came from PN junction loss, and ~1.3 dB of optical loss was from passive routing waveguides and the thermally tunable MZM optical splitter and combiner used within each device. The electro-optical (EO) S21 3 dB bandwidth for the MZM with integrated inductors was ~12 GHz with a −1V bias, and ~18 GHz with −5V bias. The EO S21 at −5V bias is shown in Fig. 2(b). The same figure also illustrates that the MZM with no integrated inductors had a significantly reduced EO S21 bandwidth of ~9 GHz with a −5 V bias. The nearly doubled EO S21 bandwidth of the MZM with integrated inductors highlights the improvement in RF/optical velocity matching obtained when inductors are included in the electrode design. The RF transmission loss through a single inductor was extracted by comparing RF transmission through MZMs with and without inductors. The RF insertion loss of each inductor was ~1dB at 10 GHz, and ~2 dB at 20 GHz. Therefore, even though the inductors add frequency dependent RF loss to the MZM electrode that negatively impacts MZM bandwidth, the data in Fig. 2(b) shows that the benefit realized from improved impedance and velocity matching significantly outweighs the RF loss penalty. Furthermore, lower loss inductors can be implemented with a slightly larger footprint, which should further improve MZM bandwidth.
Fig. 2 a) Periodic loaded MZM with no integrated inductors in the electrode design with 0.8 mm of RF/optical interaction length on each arm of the push-pull MZM (1.6 mm total), b.) Measured EO S21 in 0.8 mm per side MZM with and without inductors at −5V bias.
Figure 3(a) shows a picture of an MZM that had a 2.4 mm total PN junction RF/optic interaction length (1.2 mm per MZM arm). The RF reflection measurements from this device in Fig. 3(b) illustrate S11 < −17.5 dB out to 20 GHz (extracted effective characteristic impedance ~55 Ω at −2.5V bias), compared to similar devices without inductors which showed only < −10 dB (extracted effective characteristic impedance ~30 Ω at −2.5V bias) over the same frequency range. This result illustrates that the inclusion of inductors significantly improves impedance matching to 50 Ω.
Fig. 3 a.) Picture of an MZM with integrated inductors that had a 2.4 mm total PN junction RF/optic interaction length (1.2 mm per MZM arm). Also, a schematic highlight (red paths) of the optical waveguide path in each MZM arm are shown illustrating the asymmetric MZM design and different optical delays between each PN junction loaded section in each of the two MZM arms. b.) Measurements of the electrical S11 response from the device in Fig. 3(a) with integrated inductors, and also results are shown from a second MZM with a 2.4 mm RF/optic interaction length, but without integrated inductors.
The MZMs have an optically asymmetric design, where one MZM arm has short passive optical waveguides between each loaded electrode section, and the other MZM arm has longer passive optical waveguides between each loaded electrode section. The schematic highlights (red paths) in Fig. 3(a) illustrate the asymmetric MZM design, where the short passive waveguide sections are ~130 μm and create ~1.7 ps of delay between PN junction loaded segments, and the long passive waveguide sections are ~330 μm giving ~4.4 ps of delay between loaded segments. Therefore, in the 2.4 mm MZM (1.2 mm RF/optic interaction region in each MZM arm) the light takes ~13.3 ps longer to propagate through the longer (right side) MZM arm than it takes to transverse the shorter MZM arm (left side). The S21 electro-optic bandwidth was independently measured in each of the two MZM arms using a lightwave component analyzer and launching the RF drive signal into only one of the push-pull MZM arms at a time. Figure 4(a) shows that the MZM arm with shorter optical delays had a ~12 GHz bandwidth, whereas the MZM arm with longer optical delays had a 10 GHz bandwidth.
Fig. 4 a) Electro-optic S21 bandwidth of each MZM arm from the device shown in Fig. 3(a) with −5V reverse bias. b) Electrical S21 RF propagation loss measured in the MZM with integrated inductors shown in Fig. 3(a) with a −5V bias.
The S21 electrical transmission loss through the MZM pictured in Fig. 3(a) was directly measured using an electrical network analyzer and is plotted in Fig. 4(b). From the data in Fig. 4(b) we can predict what the maximum bandwidth of an MZM could be with co-propagating RF/optic fields, and perfect impedance and RF/optic velocity matching, since the dominant bandwidth limiting mechanism in an impedance and velocity matched MZM is electrode loss [43]. For example, a traveling wave MZM that is perfectly velocity and impedance matched has a 3 dBe EO S21 bandwidth response when the peak-to-peak RF drive voltage, averaged over the entire length of the MZM electrode, is 70.7% of the maximum averaged RF drive over the frequency range of measurement [42, 43]. In a standard traveling wave MZM the 3 dB-electrical EO S21 response condition is realized when the RF power is reduced by −6.4 dB at the end of the MZM electrode [43]. Therefore, the frequency at which a −6.4 dB RF power loss is realized in the MZM electrode represents the maximum possible 3 dB-electrical EO S21 bandwidth in a co-propagating MZM, when perfect velocity and impedance matching is present. The data in Fig. 4(b) shows that the −6.4 dB RF electrical power loss happens at an RF frequency of ~20 GHz. So for a standard traveling wave MZM with perfect velocity and impedance matching a 20 GHz 3 dB-electrical EO S21 bandwidth would be expected. However, since the measured bandwidth of our device gives a maximum value of 12 GHz, there is some bandwidth limitation in addition to that from the electrode loss. The excellent impedance matching demonstrated by this device in the RF S11 data of Fig. 3(b) suggests this additional bandwidth limitation is due to RF/optic velocity mismatch within the MZM.
To better understand RF/optical velocity matching through the device, the optical time of flight was measured using a LUNA 5013 Optical Vector Analyzer operated in the optical frequency domain reflectometer mode. Optical time of flight measurements were made on the MZMs shown in Fig. 1(c) and Fig. 3(a), which have different RF/optical interaction lengths but nominally identical routing waveguides outside of the MZMs. The time of flight through the optical waveguide sections highlighted with solid red arrows within the MZM segment pictured in Fig. 1(d) was then determined by subtracting the optical time of flight through the MZM in Fig. 1(c) from that through the MZM in Fig. 3(a). Using this technique the optical time of flight to travel through the section shown in Fig. 1(d) was measured to be ~14.1 ps. Using a lightwave component analyzer the RF time of flight through the same MZM segment was similarly measured to be ~11.3 ps at 2 GHz (RF effective group index (nRF) ~6.4) and ~8.6 ps at 18 GHz (nRF ~5) with inductors, and ~7.6 ps (nRF ~4.4) and ~5 ps (nRF ~3) at 2 GHz and 18 GHz, respectively, without inductors. This result shows the inductors significantly improve RF/optic velocity matching in the MZM with the optical ‘double pass’ configuration. Since at 18 GHz there is a |8.6 ps - 14.1 ps| = 5.5 ps RF/optic velocity mismatch in each ‘Fig. 1(d) segment’ in the MZM with inductors, and there are 3 such segments along the length of the 2.4 mm MZM, there is a total of ~16.5 ps of velocity mismatch along the length of the 2.4 mm MZM with inductors. For comparison, we calculate that for a ‘standard’ TW MZM with co-propagating RF/optic fields and a similar electrode loss and RF/optic velocity mismatch, the expected bandwidth would be ~12.8 GHz. This indicates that the counter propagating RF/optic interaction segments in the MZM design studied here did not have a significantly negative impact on device bandwidth, and that the RF/optic interaction regions were appropriately designed to provide a lumped element response within the MZM architecture.
Figure 5(a) shows a 3.1 dB extinction ratio (ER) 25 Gb/s eye diagram attained with 1.55 μm light, while driving the 1.6 mm MZM (0.8 mm interaction length in each MZM arm) with inductors using a 1.75 Vpp RF drive on each arm of the MZM, and a DC bias of −0.8 V. A longer 2.4 mm (1.2 mm per MZM arm) push pull MZM was driven with 5 Vpp and −2.5 V DC bias, producing a 25 Gb/s eye diagram having a >11 dB extinction ratio as shown in Fig. 5(b). The eye shown in Fig. 5(c) is obtained with identical drive conditions to that used in Fig. 5(b), except that the optical and RF signals were coupled into the device in opposite directions. The resulting closed eye diagram confirms that the distributed-electrode MZM device does in fact operate as a travelling-wave device, and illustrates the positive impact the distributed-electrode design has on performance.
Fig. 5 a) 3.1 dB ER 25 Gb/s eye with 1.75 Vpp drive and −0.8 V DC bias on each MZM arm from the device shown in Fig. 1(c), 1(b)) 11.2 dB ER eye from 5 Vpp drive and −2.5 V DC bias on each MZM arm from the device shown in Fig. 3(a), 3(c)) Same operating conditions as 5b but optical and RF inputs are launched in counter-propagating directions.
4. Conclusion
We have demonstrated a novel travelling-wave lumped-element distributed electrode MZM design based on the utilization of integrated RF inductors with reverse-biased double-pass highly-capacitive phase shifters. Using the double pass optical design within the RF-optical interaction region can significantly reduce the MZM footprint. Our results show that the effective characteristic impedance of the MZM electrode with integrated inductors was >50 Ω, and that the RF time of flight was faster than the optical time of flight through the MZM. The time delay of the RF signal is given by , and the impedance, , where L is inductance and C is capacitance. Therefore, the MZM electrode RF time delay could be increased, and the characteristic impedance decreased. by simply increasing the capacitance of the of the PN-junction phase shifters, which could be achieved by use a higher doping level in the plasma-dispersion reversed-biased phase shifters. However, increasing the phase shifter doping will not only increase the MZM electrode line capacitance, but will also have the additional effect of increasing optical loss and decreasing Vπ·L. We also note that if a larger change in RF time delay were required than could be realized by simply increasing the phase shifter doping, then both the capacitance and inductance could be increased simultaneously, as long as the increase in capacitance was larger than the increase in inductance. This would achieve a larger increase in RF time delay for given decrease in impedance.
Monolithically integrating inductors into the MZM design is a capability unique to CMOS platforms and enables ultra-low-Vπ·L phase shifters to be used, which tend to have high line capacitance. Designs presented here were not optimized for minimal footprint. However, an optimized double pass waveguide DE MZM design can potentially reduce MZM footprint by more than a factor of two, allowing low drive voltages and broadband operation in a relatively compact and temperature insensitive design.
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