All-optical and broadband microwave fundamental/sub-harmonic I/Q down-converters

Yongsheng Gao; Aijun Wen; Wei Jiang; Yangyu Fan; You He

doi:10.1364/OE.26.007336

1. Introduction

Microwave in-phase/quadrature (I/Q) mixers, including up-converter and down-converter, are essential modules in modern electronic systems. For example, a microwave I/Q down-converter is the kernel module of a zero immediate frequency (zero-IF) receiver [1], an image-reject receiver [2], a Doppler frequency shift estimation system [3], or a microwave phase/frequency discriminator [4]. As shown in Fig. 1, a conventional microwave I/Q mixer contains two frequency down-conversion channels with 90 degree phase difference, which is often realized by an in-phase divider, a hybrid LO divider and two mixers. When it is used for down-conversion, the coming radio frequency (RF) signal can be down-converted by the local oscillator (LO) signal into I/Q parts (IF or baseband signals). However, limited by frequency-dependent electronic dividers and mixers, the commercial microwave I/Q mixer suffers serious I/Q imbalance in wideband applications and finally faces I/Q aliasing or image interference problems [5].

Fig. 1 Conventional microwave I/Q down-converter.

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Photonic-based microwave mixers and phase shifters have attracted much attention and research in recent years [6–12], mainly due to the merits of large bandwidth, low frequency-dependent loss, and electromagnetic immunity [13–16]. The integration of photonic microwave mixer and phase shifter have been also studied to realize phased array in a superheterodyne transceiver [17, 18] or phase detection in a phase noise analyzer [19, 20]. Combining the microwave mixer and phase shifter in the optical domain, photonic I/Q down-converters are consequently reported in recent years. The quadrature phase difference between the two frequency down-conversion channels can be achieved using dispersion-induced phase shift, which is verified in the photonic vector demodulator proposed in [21] and the radar detector in [22]. The quadrature phase difference can be also realized through differentiated RF/LO driving and proper bias control for the Mach-Zehnder modulator [23, 24]. However, due to frequency dependence of fiber dispersion and electronic couplers, the photonic I/Q down-converter is limited in the operating frequency and signal bandwidth. In [25, 26], an optical hybrid coupler is used to mix RF/LO signals and build the I/Q down-conversion channels. However, the discrete parallel links suffer serious environmental disturbance. This problem may be mitigated by minimize the path separation [27], but not eliminated. The polarization division multiplexing technique can solve this problem, as described in the photonic I/Q down-converter based on tandem phase modulation and polarization modulation [28]. However, the tandem photonic converter suffers more optical loss and the conversion gain is significantly lower than the parallel one [29]. Recently, we proposed a photonic microwave I/Q down-converter using a polarization division multiplexing Mach-Zehnder modulator (PDM-MZM) [30]. However, this system suffers large dc offset and second-order intermodulation distortion (IMD2) due to the self-beating after photodetection. In addition, only fundamental microwave I/Q down-conversion can be obtained using the PDM-MZM. In [31], we demonstrate an all-optical microwave image-reject down-converter based on parallel modulation and wavelength division multiplexing. Then we describe a photonic microwave zero-IF receiver based on polarization and wavelength division multiplexing techniques [32].

In this work, we promote our work further and demonstrate an all-optical and broadband I/Q down-converter based on a polarization division multiplexing dual-parallel Mach-Zehnder modulator (PDM-DPMZM). Using the modulator, two parallel optical signals modulate the RF/LO signals, and are then combined into a polarization-multiplexed signal. After optical filtering, power spitting, and independent polarization adjustment, I/Q frequency down-conversion channels are finally obtained. The LO modulation index is properly selected to optimize the conversion gain and noise figure (NF). Due to the all-optical manipulation, the proposed I/Q down-converter exhibits wide operating band and good I/Q balance. In addition, the I/Q down-conversion channels are both balanced detected to suppress the even-order term, which significantly reduces the dc offset and distortions and improve the system dynamic range. Based on the proposed system, image-reject superheterodyne reception and zero-IF I/Q demodulation are demonstrated in the experiment. In addition to fundamental I/Q down-conversion, efficient sub-harmonic I/Q down-conversion is also realized, which not only lowers the frequency requirement for the LO signal, but also reduces the LO leakage.

2. Principles

The all-optical fundamental/sub-harmonic microwave I/Q down-converters are described in Fig. 2. An optical carrier, denoted by $E_{i n} (t)$ , is generated from a laser diode (LD) and injected into the PDM-DPMZM. The PDM-DPMZM contains two sub-DPMZMs (X-DPMZM and Y-DPMZM) in parallel. Both X-DPMZM and Y-DPMZM have two sub-modulators (Xa, Xb, Ya and Yb). The Xa in X-DPMZM is driven by the RF signal while the Xb is left open. Symmetrically, the Ya in Y-DPMZM is driven by the LO signal while the Yb is left open. Xa, Xb, Ya and Yb are all biased at the null points to suppress the optical carriers.

Fig. 2 Schematic diagram of proposed all-optical fundamental/sub-harmonic microwave I/Q down-converters.

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Assuming the RF signal is small and is expressed as $V_{R} \sin (Ω_{R} t)$ , where $V_{R}$ is the amplitude and $Ω_{R}$ is the angular frequency, the optical field after the sub-modulator Xa is

\begin{matrix} E_{X a} (t) = E_{i n} (t) {\exp [j m_{R} \sin (Ω_{R} t)] - \exp [- j m_{R} \sin (Ω_{R} t)]} / 4 \\ = E_{i n} (t) {\sum_{n = - \infty}^{\infty} J_{n} (m_{R}) \exp (j n Ω_{R} t) [1 - {(- 1)}^{n}]} / 4 \\ \approx E_{i n} (t) J_{1} (m_{R}) [\exp (j Ω_{R} t) - \exp (- j Ω_{R} t)] / 2 \end{matrix}

where

m_{R} = π V_{R} / V_{π}

denotes the RF modulation index, and V_π is the half-wave voltage of the modulator.

J_{n} (\cdot)

is the n-th order Bessel function of first kind, and n is an integer. We can see that the optical signal contains RF modulated upper and lower sidebands, whose spectrum is shown in Fig. 2. As the sub-modulator Xb is biased at the null point and not driven by the RF signal, there is no optical signal travelling out. Therefore, the optical field after X-DPMZM is

\begin{matrix} E_{X} (t) = E_{X a} (t) / \sqrt{2} \\ = E_{i n} (t) J_{1} (m_{R}) [\exp (j Ω_{R} t) - \exp (- j Ω_{R} t)] / 2 \sqrt{2} \end{matrix}

Similarly, assuming the LO signal is expressed as

V_{L} \sin (Ω_{L} t)

, where

V_{L}

is the amplitude and

Ω_{L}

is the angular frequency, the optical filed after Y-DPMZM is

\begin{matrix} E_{Y} (t) = E_{i n} (t) {\exp [j m_{L} \sin (Ω_{L} t)] - \exp [- j m_{L} \sin (Ω_{L} t)]} / 4 \sqrt{2} \\ = E_{i n} (t) {\sum_{n = - \infty}^{\infty} J_{n} (m_{L}) \exp (j n Ω_{L} t) [1 - {(- 1)}^{n}]} / 4 \sqrt{2} \\ \approx E_{i n} (t) J_{1} (m_{L}) [\exp (j Ω_{L} t) - \exp (- j Ω_{L} t)] / 2 \sqrt{2} \end{matrix}

where

m_{L} = π V_{L} / V_{π}

denotes the LO modulation index. This optical signal travels through a 90 degree polarization rotator (PR). Then the optical signals from the two sub-DPMZMs are combined by the followed polarization beam combiner (PBC), and a polarization-multiplexed optical signal expressed below is generated after the PDM-DPMZM

E_{P D M} (t) = | \begin{array}{l} E_{X} \\ E_{Y} \end{array} | = E_{i n} (t) \cdot | \begin{array}{l} J_{1} (m_{R}) [\exp (j Ω_{R} t) - \exp (- j Ω_{R} t)] \\ J_{1} (m_{L}) [\exp (j Ω_{L} t) - \exp (- j Ω_{L} t)] \end{array} | / 2 \sqrt{2}

The optical signal is amplified by an erbium doped fiber amplifier (EDFA) and filtered by an optical bandpass filter (OBPF). The OBPF is used to get one sideband of the optical signal. The polarization-multiplexed single sideband optical signal after the OBPF is

E_{O B P F} (t) = E_{i n} (t) \cdot | \begin{array}{l} J_{1} (m_{R}) \exp (j Ω_{R} t) \\ J_{1} (m_{L}) \exp (j Ω_{L} t) \end{array} | / 2 \sqrt{2}

The corresponding spectrum is also plotted in Fig. 2. The optical signal is then divided into two channels (I/Q) by an optical splitter. In each channel, a polarization controller (PC) and a polarization beam splitter (PBS) is followed to form two optical signals:

E_{1} (t) = E_{i n} (t) [J_{1} (m_{R}) \exp (j Ω_{R} t) \cos α + J_{1} (m_{L}) \exp (j Ω_{L} t) \sin α \exp (j φ)] / 4

E_{2} (t) = E_{i n} (t) [J_{1} (m_{R}) \exp (j Ω_{R} t) \sin α - J_{1} (m_{L}) \exp (j Ω_{L} t) \cos α \exp (j φ)] / 4

where

α

is the polarization azimuth and

φ

is the phase difference between the two polarization components. In order to realize balanced detection, the PC is adjusted to set

α = 45^{0}

. In the balanced photodiode (BPD), the two photocurrents subtract each other, and the following current is generated after the BPD:

i (t) \propto {| E_{1} (t) |}^{2} - {| E_{2} (t) |}^{2} \propto J_{1} (m_{R}) J_{1} (m_{L}) \cos [(Ω_{R} - Ω_{L}) t - φ]

After balanced detection, the desired difference frequency term, or the IF signal, is obtained and the undesired dc offset and even-order distortions are cancelled out. We can also find that the phase of the IF signal is related to $φ$ , which can be arbitrarily tuned through polarization control. In order to build I/Q frequency down-conversion channels, the PCs in the two channels should be independently adjusted as follows.

In the first channel, the PC is adjusted to set $φ = 0^{0}$ , and the two spectra after the PBS are shown in Fig. 2. The photocurrent after BPD can be written as

i_{I} (t) \propto J_{1} (m_{R}) J_{1} (m_{L}) \cos [(Ω_{R} - Ω_{L}) t]

So an in-phase down-converted IF signal is obtained. In the other channel, the PC is adjusted to set

φ = 90^{0}

, and the two spectra after the PBS are also shown in Fig. 2. The photocurrent after BPD can be written as

i_{Q} (t) \propto J_{1} (m_{R}) J_{1} (m_{L}) \sin [(Ω_{R} - Ω_{L}) t]

So a quadrature down-converted IF signal is obtained. Finally, the proposed system becomes a microwave I/Q down-converter.

In an RF receiver with a superheterodyne architecture, the two IF signals can be combined by a hybrid coupler for image rejection [30]. In an RF receiver with a zero-IF architecture, the LO frequency is equal to the RF carrier frequency, and the I/Q baseband signals are obtained after I/Q down-conversion.

This system can also provide sub-harmonic I/Q down-conversion. In this operation mode, a relative large LO modulation index is applied and the Ya in Y-DPMZM is biased at the peak point to generate the second-order sidebands, along with the optical carrier. The Yb in Y-DPMZM is biased at a specific point, whose bias phase shift is denoted by $θ$ . The main modulator Y-DPMZM is biased at the null point. The optical filed after Y-DPMZM is rewritten as

\begin{matrix} E_{Y} (t) = E_{i n} (t) {\exp [j m_{L} \sin (Ω_{L} t)] + \exp [- j m_{L} \sin (Ω_{L} t)] - 2 \cos (θ / 2)} / 4 \sqrt{2} \\ = E_{i n} (t) {\sum_{n = - \infty}^{\infty} J_{n} (m_{L}) \exp (j n Ω_{L} t) [1 + {(- 1)}^{n}] - 2 \cos (θ / 2)} / 4 \sqrt{2} \\ \approx E_{i n} (t) {J_{2} (m_{L}) [\exp (j 2 Ω_{L} t) + \exp (- j 2 Ω_{L} t)] + J_{0} (m_{L}) - \cos (θ / 2)} / 2 \sqrt{2} \end{matrix}

The amplitude of the optical carrier output from Yb is related to $θ$ , which can be adjusted to equate with the optical carrier output from Ya. By biasing the Y-DPMZM at the null point, the optical carriers cancel each other out. The optical carrier suppression condition can be expressed as

J_{0} (m_{L}) = \cos (θ / 2)

Then the optical field after Y-DPMZM is written as

E_{Y} (t) = E_{i n} (t) J_{2} (m_{L}) [\exp (j 2 Ω_{L} t) + \exp (- j 2 Ω_{L} t)] / 2 \sqrt{2}

The following OBPF is now used to filter one of the second-order sidebands. Keeping other system configurations unchanged, the photocurrents after balanced detection are

i_{I} (t) \propto J_{1} (m_{R}) J_{2} (m_{L}) \cos [(Ω_{R} - 2 Ω_{L}) t]

i_{Q} (t) \propto J_{1} (m_{R}) J_{2} (m_{L}) \sin [(Ω_{R} - 2 Ω_{L}) t]

Therefore, a photonic sub-harmonic microwave I/Q down-converter is obtained.

3. Experiment and results

3.1 Experimental setup

An experiment is carried out in accordance to Fig. 2. The optical carrier output from an LD (Emcore 1782) has a wavelength of about 1552.2 nm, a power of 50 mw and a relative intensity noise of below −160 dBm. The optical carrier is injected into a PDM-DPMZM (FTM7977HQA) with a linear polarization. The modulator has a half-wave voltage of about 3.5 V and an extinction ratio of 30 dB. The RF and LO signals are generated from a vector signal generator (SMW200A) and a microwave signal generator (N5183A MXG). The maximum frequencies of the two signal generators are both 40 GHz. The modulator is driven by the RF and LO signals according to Fig. 2, and its dc biases are manually controlled using two dc stabilized power supplies (E3631A). The EDFA (KPS-STD-C-19-HG) has a fixed output power of 17 dBm and an NF of 4.5 dB. The followed OBPF has a 3 dB bandwidth of 0.6 nm and a 20 dB bandwidth of 0.93 nm. The response curve of the OBPF is measured by an optical spectrum analyzer (Advantest Q8384) with a resolution of 0.01 nm, and the result is plotted in Fig. 3 using a black dot line. The two BPDs are fabricated using four 1 GHz photodiodes with a responsivity of 1.1 A/W. The matched circuits after the photodiodes are designed symmetrically to reduce length mismatching. Due to the limited experiment condition, the length matching of the two BPDs is not measured in the experiment.

Fig. 3 Response curve of OBPF, and optical spectra before and after OBPF in fundamental I/Q down-converter.

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3.2 Fundamental I/Q down-conversion

First, the fundamental I/Q down-conversion is demonstrated and measured. The four sub-modulators in the PDM-DPMZM are all biased at the null points. A 26 GHz RF signal is directly down-converted to 100-MHz IF signals by the fundamental I/Q down-converter using a 25.9 GHz LO signal. The optical spectra before and after the OBPF are plotted in Fig. 3 using a red dashed line and a blue solid line, respectively. We can see that after optical filtering, the lower first-order optical sidebands are well suppressed by about 25 dB.

By tuning the LO power from 2 to 17 dBm in 1 dB step, the conversion gain and NF are measured and plotted in Fig. 4. In this measurement, only a single photodiode is used. As the optical power injecting into the photodiode keeps unchanged (fixed output power of EDFA), a lower LO power will help optimize the power ratio between the LO and RF sidebands, and thus maximize the beating terms (or the down-converted IF signals) [7, 8, 32, 33]. Therefore, the conversion gain varies inversely with the LO power. However, as the conversion gain is improved by further pumping the EDFA, the amplified spontaneous emission noise from the EDFA is inevitably enhanced. What’s more, a low LO power will lead to a small input power of the EDFA. To ensure the desired output power, a large power gain is required, which may exceed the ability of the EDFA. This is the reason why the conversion gain increases little and the NF is obviously deteriorated with a small LO power. As a tradeoff, the LO power in the following experiment is set to 10 dBm.

Fig. 4 Measured conversion gain and NF versus LO power in fundamental I/Q down-converter.

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A two-tone test is then conducted and the spurious free dynamic range (SFDR) is measured. A two-tone RF signal (26 and 26.01 GHz) with a power of 0 dBm in each tone are down-converted to an IF signal with the fundamental frequencies of 100 and 110 MHz. The spectrum of the IF signal after one photodiode is measured by an electrical signal analyzer (FSW50) and shown in Fig. 5(a). Except for the two fundamental terms, small third-order intermodulation distortions (IMD3s) are also found at the frequencies of 90 and 120 MHz, which are due to the inherent nonlinearity of the modulator and photodiode. In addition, an IMD2 with a power of −18.9 dBm can also be observed at the frequency of 10 MHz, which originates from the self-beating of the two-tone RF signal. When both photodiodes are illuminated, balanced detection is achieved with proper polarization alignment. Figure 5(b) plots the IF spectrum after BPD. We can see the IMD2 at 10 MHz is reduced to −69 dBm, which is 50.1 dB lower than that without balanced detection. Meanwhile, the fundamental terms and IMD3s are both improved by 5.5 dB, close to the theoretical prediction (6 dB). The well suppressed IMD2 shown in Fig. 5(b) also indicates a good responsivity balance and length matching of the BPD.

Fig. 5 Down-converted signals with (a) single photodiode, and (b) BPD in fundamental I/Q down-converter.

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Then the power of the two-tone RF signal is gradually changed from −30 to 6 dBm and the down-converted IF fundamental term, the IMD2, the IMD3 and the noise floor are measured, as shown in Fig. 6. When a small RF power is applied, the IMD2 and IMD3 are lower than the noise, so the number of points of fundamental, IMD2 and IMD3 is not equal. A conversion gain of −2.4 dB and an NF of 40.6 dB are obtained. Note that the conversion gain and NF are slightly improved compared with the results without balanced detection in Fig. 4. The third-order input intercept point (IIP3) is 28.6 dBm. The second-order SFDR (SFDR₂) and third-order SFDR (SFDR₃) are 100.2 dB·Hz^1/2 and 108 dB·Hz^2/3, respectively.

Fig. 6 IF output power versus RF input power in fundamental I/Q down-converter.

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The phase and power balance are then investigated. A 26 GHz RF signal is down-converted into 100 MHz I/Q signals using 25.9 GHz LO signal. The IF waveforms are observed using a dual-channel oscilloscope (DPO7254). The waveform of the in-phase signal (as the oscilloscope trigger) is drawn using a black line in Fig. 7. Through polarization adjustment, the relative phase of the quadrature signal is continuously and arbitrarily tuned. The polarization state (φ and α) is adjusted according to the down-converted I/Q waveforms. When α = 45 degrees, the dc offset after the BPD is eliminated. When φ is 0/90 degrees, the phase difference between the two IF waveforms are quadrature. As an example, the waveform of the quadrature IF signal with relative phase of 0, 90, 180 and 270 degrees are captured and plotted in Fig. 7. We can also find that the amplitudes of the waveforms with different phases are highly consistent.

Fig. 7 Waveforms of in-phase IF signal and quadrature IF signal with relative phase of 0, 90, 180, 270 degrees in fundamental I/Q down-converter.

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In the followed experiment, the phase difference between the two IF signals is set to 90 degrees. The RF frequency is changed from 5 to 40 GHz in 1 GHz step. By changing the LO frequency accordingly, the IF frequency remains at 100 MHz. The powers of the I/Q signals and their phase difference are measured and shown in Fig. 8(a). We can see the 3 dB RF operating bandwidth ranges from 7 to 40 GHz. Over this large bandwidth, the maximum power imbalance is 0.5 dB and the maximum phase imbalance is only 0.9 degrees. Then the RF frequency is set to 26 GHz. By gradually tuning the LO frequency, the IF frequency is changed from 0.1 to 1 GHz in 0.1 GHz step. As shown in Fig. 8(b), the power fluctuation over this operating bandwidth is 3.1 dB and the maximum power imbalance is 0.7 dB. The phase imbalance remains below 0.8 degrees.

Fig. 8 Measured I/Q phase and power imbalance versus (a) RF frequency and (b) IF frequency in fundamental I/Q down-converter.

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3.3 Fundamental image-reject receiver

When the proposed I/Q down-converter is applied for an image-reject receiver, the I/Q IF signals will be combined using an analog or digital 90 degree hybrid coupler to eliminate the image product. A 26 GHz RF signal is down-converted to two 500 MHz IF signals using a 25.5 GHz LO signal. The I/Q IF signals are combined by an analog hybrid coupler and the coupled signal is sent to the oscilloscope and signal analyzer in turn. The waveform and spectrum of the desired 500 MHz signal are plotted using blue solid lines in Figs. 9(a) and 9(b), respectively. Next, the RF power is fixed and its frequency is changed to 25 GHz to act as the image interference. The waveform and spectrum captured by the instruments are drawn using red dashed lines in Figs. 9(a) and 9(b), respectively. A weak signal at 500 MHz, which is the image product, is observed. As compared with the desired IF signal, the image product is suppressed by 49.5 dB.

Fig. 9 Desired signal and image product after image-reject down-conversion based on fundamental I/Q down-converter: (a) waveforms and (b) spectra. LO frequency: 25.5 GHz; desired RF frequency: 26 GHz; image RF frequency: 25 GHz.

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3.4 Fundamental zero-IF receiver

The proposed I/Q down-converter can be also configured as a zero-IF receiver. In the experimental demonstration, a 26 GHz RF vector signal with 100 MBaud 16 quadrature amplitude modulation (16QAM) is directly down-converted to baseband using a synchronized 26 GHz LO signal. The I/Q baseband signals are simultaneously analyzed to calculate the error vector magnitudes (EVMs). The power of the RF vector signal is tuned from −31 to 10 dBm, and the EVM curve is plotted using a blue line in Fig. 10. The EVM remains below 10% with the RF power sweeping from −25 to 7 dBm. As examples, the constellation diagrams with RF powers of −25 and 5 dBm are shown in the insets, and the calculated EVMs are 8.9% and 8%, respectively. Next, the RF carrier frequency is changed to 36 GHz then direct-converted using a 36 GHz LO signal. The EVM curve is drawn in Fig. 10 using a red line. We can see the two EVM curves with different RF carrier frequencies are quite similar, which indicates a good frequency tunability of the zero-IF receiver.

Fig. 10 Demodulated EVM versus RF input power with 26 and 36 GHz carrier frequencies in fundamental zero-IF receiver. Insets: constellation diagrams. Modulation format: 16QAM; signal bandwidth: 100 MBand.

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The LO/RF isolation as a function of operating frequency is also tested and plotted in Fig. 11. It keeps below −30 dB over 0–40 GHz and below −40 dB over 0–20 GHz. The RF and LO ports are electro-optic shielded, and the residual leakage mainly results from the crosstalk between the two drive electrodes in the modulator. Nevertheless, this result is much better than conventional microwave I/Q mixers. In some special applications, a much higher LO/RF isolation may be required to reduce the LO leakage, then the followed sub-harmonic I/Q down-conversion may be more advantageous.

Fig. 11 Measured LO/RF isolation versus operating frequency.

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3.5 Sub-harmonic I/Q down-conversion and zero-IF receiving

Next, the system is reconfigured as a sub-harmonic I/Q down-converter. A 36 GHz RF signal is converted to a 100 MHz IF signal by a 17.95 GHz LO signal. The dc biases in X-DPMZM remain unchanged. The Ya modulator in Y-DPMZM is operated at the peak bias point to suppress first-order optical sidebands and enhance second-order ones. According to the Bessel functions, the modulation efficiency of second-order sidebands is lower than the first-order ones. So in order to improve the sub-harmonic conversion gain, a relative large LO power of 18 dBm is selected in the experiment. The dc biases of the sub-modulator Yb and the main modulator Y-DPMZM are tuned according to the analysis to suppress the optical carrier. Figure 12 shows the optical spectra before and after optical filtering. The upper second-order sideband becomes the dominant term and the harmonic suppression ratio is about 18 dB. The residual first- and third-order LO sidebands result from the limited modulator extinction ratio.

Fig. 12 Filter response and optical spectra before and after OBPF in sub-harmonic I/Q down-converter.

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Similar to the fundamental I/Q down-converter, the phase difference between the two IF signals can be arbitrarily adjusted through the PCs. The waveforms of the in-phase IF signal and the quadrature IF signal with different relative phases are drawn in Fig. 13. The phase difference between the two waveforms can be adjusted to the desired value while their amplitudes remain unchanged.

Fig. 13 Waveforms of in-phase IF signal and quadrature IF signal with relative phase of 0, 90, 180, 270 degrees in sub-harmonic I/Q down-converter.

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The measured I/Q phase and power imbalance are shown in Fig. 14. In the 3 dB RF operating bandwidth (10–40 GHz), the maximum power imbalance is 0.6 dB and the phase imbalance is below 1 degree. With the IF frequency from 0.1 to 1 GHz, the IF signals have a maximum power fluctuation of 3.3 dB. The maximum power imbalance is 0.5 dB and the phase imbalance is below 0.8 degrees.

Fig. 14 Measured I/Q phase and power imbalance versus (a) RF frequency and (b) IF frequency in sub-harmonic I/Q down-converter.

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The sub-harmonic I/Q down-converter is easy to be configured as an image-reject receiver with results similar to those demonstrated in the fundamental one. Next, only the zero-IF receiver is demonstrated using the sub-harmonic I/Q down-converter. A 36 GHz RF vector signal with a modulation format of 16QAM and a bandwidth of 100 MBand is direct-converted to I/Q baseband signals using an 18 GHz LO signal. The measured EVM with different RF power is plotted in Fig. 15. The EVM is below 10% with a power range of 31 dB (−27 to 4 dBm). Two constellation diagrams with RF powers of −25 and 5 dBm are also shown in Fig. 15, with the calculated EVMs of 8.8% and 10.5%, respectively.

Fig. 15 Demodulated EVM versus RF input power in sub-harmonic zero-IF receiver. Insets: constellation diagrams. Modulation format: 16QAM; signal bandwidth: 100 Mband; RF frequency: 36 GHz; LO frequency: 18 GHz.

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4. Discussion

Although the photonic techniques exhibit superiority in bandwidth, loss and electromagnetic isolation, how it can be efficiently applied in modern electronic systems still remains challengeable. Much work has proved the feasibility of the photonic-based single modules such as photonic-based microwave mixer, filter, phase shifter, frequency multiplier, etc. However, practical electronic systems, like an RF transmitter/receiver or microwave instrument, are often complex systems that involve several modules mentioned above. Due to the low efficiency in electro-optic modulation and photodetection, direct-integration of these modules will often lead to the accumulation of noise and nonlinearity, and finally cause performance degradation. The main work in this paper is to design a multifunctional but simple microwave photonic system that is closer to practical application. In the proposed fundamental I/Q down-converter, the microwave down-conversion and phase shifting are efficiently integrated in a simple and all-optical system. Additional frequency doubling for the LO signal is also introduced to the sub-harmonic I/Q down-converter. The highlight of the proposed system is that both the frequency down-conversion and phase shifting obtain a good performance in the experiment: high conversion gain, high IIP3 and large SFDR in frequency down-conversion; arbitrary and continuous tuning in phase shifting.

4.1 Observations

Thanks to the all-optical configuration, the proposed system exhibits an ultra-wide bandwidth: 7–40 GHz in the fundamental I/Q down-converter and 10–40 GHz in the sub-harmonic one. Furthermore, due to the frequency-independent and power-independent phase shifting, ultra-low I/Q phase imbalance (<1 degree) and power imbalance (<0.7 dB) are achieved over the wide operating bandwidth. When applied in an image-reject receiver, the excellent I/Q balance provides a large image rejection; while in a zero-IF receiver, it can avoid the aliasing of demodulated I/Q baseband signals. The wideband feature makes the I/Q down-converter versatile in much of modern microwave and millimeter-wave applications. For example, it may be attractive in a frequency-agile radar [22] or multi-band satellite payload.

The quadrature frequency down-conversion channel in the proposed system can also be used as a phase or frequency discriminator, which can be applied in a phase-locked loop [34], a phase noise analyzer [19], an angle-of-arrival measurement system [35], and so forth. A typical application of the frequency discriminator is Doppler frequency shift estimation [3]. The transmitted signal acts as the LO signal and the echo signal with Doppler frequency shift acts as the RF signal. The numerical value of the Doppler frequency shift is equal to the frequency of the two down-converted IF signals, and the direction of the Doppler frequency shift can be derived from the phase difference between the two IF signals.

The dc offset and even-order distortion may be not concerned in an image-reject receiver. Nevertheless, in a zero-IF receiver, the dc offset and even-order (mainly second-order) distortion will fall into the down-converted I/Q baseband and cannot be filtered out. Similar problem also exists in a phase/frequency discriminator [36]. In the proposed fundamental and sub-harmonic I/Q down-converters, the dc and second-order distortion are eliminated by balanced detection, and thus a high SFDR₂ is obtained.

The LO leakage, caused by limited LO/RF isolation, is another important problem that limits the practical application of electronic zero-IF receivers. On one hand, the LO leakage will travel out through the receiver antenna and become a strong interference for other channels. On the other hand, the LO leakage may reflect back, mix with the LO signal and finally generate new dc offset [37]. Unfortunately, this dc offset cannot be eliminated by balanced detection. Harmonic frequency conversion is an efficient solution for LO leakage [5]. Photonic harmonic I/Q down-conversion has been reported in [12], with the help of electrical phase shifting techniques. However, the LO modulated optical signal contains many undesired components, which not only reduces the conversion gain, but also aggravates the dc offset induced by LO self-mixing. In the proposed sub-harmonic I/Q down-converter, the spectral purity of the optical signal is greatly improved, as shown in Fig. 13. As a consequence, the dc offset is not obvious, which is verified by the IF waveforms in Fig. 7 and Fig. 14, and the demodulated constellation diagrams in Fig. 11 and Fig. 16. Last but not the least, the proposed sub-harmonic I/Q down-converter will reduce half of the bandwidth requirements for the LO signal and the driver amplifier.

Fig. 16 Designed polarization-demultiplexed I/Q photodetector.

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4.2 Challenges

Some challenges also exit in the current system yet. First, the modulator bias drift is a problem. In the fundamental I/Q down-converter, the four sub-modulators are all biased at the null points, which will be easily controlled by an automatic feedback circuit [38, 39]. However, when the system is operated in the sub-harmonic I/Q down-conversion, the sub-DPMZM driven by the LO signal is biased at specified points expressed in (10). As far as we know, the automatic bias control for DPMZM with specified points is not commercially available.

Second, the polarization is sensitive to the environment. As the I/Q phase difference and balanced detection are directly dependent on the polarization, the polarization jitter will significantly cause performance degradation. In the experiment, the polarization is manually tuned, which is not feasible in practical applications. A solution is to apply automatic polarization control [40], but may increase the system complexity.

4.3 Future plan

To improve the system stability and reduce the complexity, our future plan is to fabricate a polarization-demultiplexed I/Q photodetector to replace the optical splitter, PCs and PBSs in Fig. 2. A possible polarization-demultiplexed I/Q photodetector is drawn in Fig. 16, including three PBSs, a half waveplate, a quarter waveplate, a 50/50 beam splitter and two BPDs. Assume the input polarization multiplexed optical signal contains two orthogonal components, denoted as X and Y. The first PBS separates X and Y components. Then the followed half/quarter waveplates, 50/50 beam splitter and two PBSs form a 2 × 4 quadrature optical hybrid, which combines the X and Y components and generates four optical signals with a 90-degree phase difference for balanced detection.

The polarization multiplexed optical after optical filtering in Fig. 2 can be expressed as

| \begin{array}{l} X = \frac{E_{i n} (t)}{2 \sqrt{2}} J_{1} (m_{R}) \exp (j Ω_{R} t) \\ Y = \frac{E_{i n} (t)}{2 \sqrt{2}} J_{1} (m_{L}) \exp (j Ω_{L} t) \end{array} |

If it is injected into this polarization demultiplexed I/Q photodetector, we can calculate that the desired I/Q signals expressed as (12) and (13) can be obtained after balanced detection. The designed polarization-demultiplexed I/Q photodetector is easily integrated in a single chip. Meanwhile, a polarization maintaining EDFA and OBPF are necessary to ensure a stable polarization in the whole system.

5. Summary

In summary, photonic microwave fundamental and sub-harmonic I/Q down-converters are proposed and experimentally measured. The system is operated in all-optical methods without any frequency-dependent modules, so a wide operating band (7–40 GHz in the fundamental I/Q down-converter, and 10–40 GHz in the sub-harmonic I/Q down-converter) and an excellent I/Q balance (maximum 0.7 dB power imbalance and 1 degree phase imbalance) are achieved. After LO power optimization and balanced detection, the conversion gain and NF are significantly improved, and the SFDR₂ and SFDR₃ at 26 GHz reach 100.2 dB·Hz^1/2 and 108 dB·Hz^2/3, respectively. When the system is used for a superheterodyne receiver, a high rejection ratio (49.5 dB at 26 GHz) is demonstrated. When it is used for a zero-IF receiver, 100 MBaud 16QAM signals at 26/36 GHz carrier frequency are well demodulated, and the EVMs keep below 10% with an RF power range of above 30 dB. Due to the broadband feature, the proposed all-optical and broadband microwave fundamental/sub-harmonic I/Q down-converter may find potential applications in the receivers of multi-band satellite, ultra-wideband radar and frequency-agile electronic warfare systems. Except for the image-reject superheterodyne receiver and zero-IF receiver, the proposed system may be also used for phase locking, phase noise analysis, angle-of-arrival measurement, Doppler frequency shift estimation, and so forth.

Funding

National Natural Science Foundation of China (NSFC) (61701412); China Postdoctoral Science Foundation (BX201700197, 2017M623238); Fundamental Research Funds for the Central Universities (G2017KY0301); National Advanced Research Foundation of China (614241105010717).

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All-optical and broadband microwave fundamental/sub-harmonic I/Q down-converters

Abstract

1. Introduction

2. Principles

3. Experiment and results

3.1 Experimental setup

3.2 Fundamental I/Q down-conversion

3.3 Fundamental image-reject receiver

3.4 Fundamental zero-IF receiver

3.5 Sub-harmonic I/Q down-conversion and zero-IF receiving

4. Discussion

4.1 Observations

4.2 Challenges

4.3 Future plan

5. Summary

Funding

References and links

Cited By

Figures (16)

Equations (16)

Optics Express