Simultaneous detection of the distance and direction for a noncooperative target based on the microwave photonic radar

Yunxin Wang; Yunxin Wang; Xueyuan Hou; Tao Li; Ziang He; Dayong Wang; Dayong Wang; Feng Yang; Feng Yang; Tao Zhou; Lu Rong; Lu Rong

doi:10.1364/OE.439486

1. Introduction

Radar, as an all-time and all-weather detection method, has been widely applied to long-range and wide-area target searching, tracking, imaging and identification [1]. With higher requirements in both military and civilian fields, excellent performance has been put forward for modern radar systems, including better resolution, real-time performance and multifunctionality. As a result, a large instantaneous bandwidth, fast data processing speed and high integration are urgently needed. However, limited by electronic bottlenecks, conventional radar systems have difficulty in satisfying the above requirements. Microwave photonic technology processes microwave signals in the optical domain [2], which has the advantages of a large bandwidth, low loss, light weight, small size and resistance to electromagnetic interference [2–7]. Recently, radar systems have achieved great progress through the combination of microwave photonics and radar technology [8–17].

To obtain the spatial location for a noncooperative target, the radar system has to detect not only the distance information but also the direction information. However, among the existing methods, distance and direction measurement functions are usually implemented by separate microwave photonic radar systems.

Several approaches have been developed to realize distance detection for a target. A mode-locked laser is used to generate a tunable linear frequency modulation (LFM) signal at the transmitter, and photon-assisted ultrafast sampling of the echo signal is applied at the receiver [8]. This method has superiority in low phase noise, a large dynamic range, and coherent transmission and reception. Nevertheless, the instantaneous bandwidth is limited by the repetition frequency of the mode-locked laser. To solve this problem, an all-optical tunable broadband LFM signal can be produced by frequency-time mapping using the optical wave dispersion effect [15]. In addition, the instantaneous bandwidth of the LFM signal can also be increased by frequency multiplication at the transmitter [16,17]. At the receiver, the echo signal is dechirped by a reference signal to reduce the sampling rate for the subsequent ADC.

The direction of the target can be measured via the angle of arrival (AOA) of the echo signal. For a single-tone echo signal, the AOA can be obtained by detecting the phase information of the downconverted signals with an oscilloscope [18,19] or by measuring the power information associated with the phase difference [20,21]. The distance of two antennas has to be smaller than half the wavelength of the received signal to avoid the angle ambiguity. The former method is relatively complex in structure. The latter is much simpler as it involves the usage of a power meter or a multimeter; however, it can only be used to detect a target in a single direction and cannot distinguish between the positive and negative directions. To achieve a larger measurement range, a phase difference is introduced to the dual parallel paths for identifying the positive or negative direction of the AOA [22,23]. Nevertheless, the accuracy is easily affected by the unstable power of the laser and the bias point drift of the modulator [24]. With regard to the broadband echo signal, a notch filter has been constructed by using a time-delay device, and the AOA can be analyzed through the notch frequency [25,26]. This scheme can be used to extract the directions of multiple echo signals simultaneously. In addition, for the LFM echo signal, the AOA can be obtained by measuring the frequencies of the two-tone signal and the slope of the LFM signal [27]. Compared with the above phase-based method, the baseline length for this method can be matched more flexibly.

The above methods can only achieve single function such as distance or direction detection. Fortunately, microwave photonics links are remarkable in terms of flexibility and compatibility, and multiple functions have been attempted to integrate into a single link. A system for simultaneous radar and frequency detection has been proposed [28]. To track a moving target, Doppler frequency shift and AOA measurements have been realized simultaneously [29]. In [30], radar detection and communication functions were achieved concurrently.

In this paper, a simultaneous distance and direction detection method for the noncooperative target based on microwave photonic radar is proposed. The decoupling of the distance and direction information are realized by the dechirp process with a specially introduced fixed-length cable (FC). At the transmitter, a linear frequency modulation signal with doubled instantaneous bandwidth is generated by a Mach-Zehnder modulator (MZM), and the reference optical signal is provided and injected into a dual-drive Mach-Zehnder modulator (DDMZM). At the receiver, the echo signals reflected from the noncooperative target are received by two antennas (RA1 and RA2) with a certain baseline length. The FC inserted after RA1 helps to effectively solve the distance and direction and distinguish between the positive and negative directions of a target. The received signal after FC is applied to drive the upper sub-PM of the DDMZM. The echo signal received by RA2 is directly injected into the lower sub-PM of the DDMZM. By analyzing the frequency relationship between the echo modulation signal and reference signal, the distance and direction of a noncooperative target can be resolved simultaneously. The positive and negative directions can also be distinguished to ensure a large angle range.

2. Principle

Figure 1 shows the schematic diagram of the proposed simultaneous distance and direction detection method for the noncooperative target. The optical carrier obtained from a narrow linewidth laser diode (LD) is injected into a Mach-Zehnder modulator (MZM). Then, the signal is modulated by an intermediate frequency linear frequency modulation signal (IF-LFM) generated from an arbitrary waveform generator (AWG). To suppress the carrier and even-order sidebands, MZM is biased at the null point. The output signal of the MZM can be expressed as

(1)$${E_{MZM}}(t) \propto rect(\frac{t}{T}) \cdot \exp (j{\omega _c}t)\left\{ \begin{array}{l} {J_1}(m)\exp [j(2\pi {f_0}t + \pi k{t^2})]\\ + {J_1}(m)\exp [ - j(2\pi {f_0}t + \pi k{t^2})] \end{array} \right\}$$

where ${\omega _c}$ is the angular frequency of the light wave from the LD; T, ${f_0}$ and k are the duration, initial frequency and chirp rate of the IF-LFM, respectively; ${J_1}(m)$ is the 1st-order Bessel function of the first kind, where $m = \pi {V_{IF - LFM}}/{V_{\pi 1}}$ is the modulation index of the MZM, in which ${V_{IF - LFM}}$ is the amplitude of the IF-LFM and ${V_{\pi 1}}$ is the half-wave voltage of the MZM.

Fig. 1. Schematic diagram of the proposed simultaneous distance and direction detection method for a noncooperative target. LD: laser diode; MZM: Mach-Zehnder modulator; AWG: arbitrary waveform generator; OC: optical coupler; PD: photodetector; EA: electrical amplifier; TA: transmit antenna; RA: receive antenna; LNA: low-noise amplifier; FC: a fixed-length cable; DDMZM: dual-drive Mach-Zehnder modulator; LPF: low-pass filter; DSO: digital storage oscilloscope.

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Then, the optical signal is split into two arms by an optical coupler (OC). One arm is sent to a photodetector (PD1) to realize IF-LFM doubling, and the output signal can be expressed as

(2)$${S_T}(t) \propto rect(\frac{t}{T}) \cdot \cos (4\pi {f_0}t + 2\pi k{t^2})$$

It can be seen that both the center frequency and bandwidth are doubled. This signal is amplified by a wideband electrical amplifier (EA) and then emitted by a transmit antenna (TA) as a detection signal.

At the receiver, the optical signal of the other arm is input to a dual-drive Mach-Zehnder modulator as a reference optical signal. The Echo signal reflected from the noncooperative target are detected by RA1 and RA2 with a baseline length of d. To distinguish between the positive and negative directions of the noncooperative target, the signal received by RA1 is transmitted via a fixed-length cable before being applied to the upper sub-PM of the DDMZM. The echo signal received by RA2 is directly fed into the lower sub-PM of the DDMZM. When the location of the noncooperative target has a certain angle of $\theta$ relative to the sightline of the two antennas, the echo signals received by the two antennas will have a certain amount of time delay, as shown in Fig. 2.

Fig. 2. The reflected signals received by two antennas for targets travelling in different directions. (a) Positive direction; (b) Negative direction.

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When the target is travelling in different directions relative to the radar sightline, the driving signals of the upper and lower sub-PMs of the DDMZM can be written as

(3)$$\left\{ \begin{array}{l} {S_{RU}}(t) \propto rect(\frac{{{t_1}}}{T}) \cdot \cos (4\pi {f_0}{t_1} + 2\pi k{t_1}^2)\\ {S_{RL}}(t) \propto rect(\frac{{{t_2}}}{T}) \cdot \cos (4\pi {f_0}{t_2} + 2\pi k{t_2}^2) \end{array} \right.$$

Because of the positive and negative directions of the noncooperative target, ${t_1}$ and ${t_2}$are different. When $0^\circ < \theta \le 90^\circ$, as shown in Fig. 2(a), ${t_1}$ and ${t_2}$ can be expressed as

(4)$$\left\{ \begin{array}{l} {t_1} = t - {\tau_R} - {\tau_{FC}}\textrm{ }\\ {t_2} = t - {\tau_R} - {\tau_\theta }\textrm{ } \end{array} \right.$$

In the case of $- 90^\circ \le \theta \le 0^\circ$, ${t_1}$ and ${t_2}$ can be denoted as

(5)$$\left\{ \begin{array}{l} {t_1} = t - {\tau_R} - {\tau_{FC}} - {\tau_\theta }\textrm{ }\\ {t_2} = t - {\tau_R}\textrm{ } \end{array} \right.$$

where ${\tau _R} = 2R/c$ is the time delay induced by round-trip propagation from the transmitter to the target, R is the distance between the target and radar system, c is the velocity of light in vacuum, ${\tau _{FC}} = {n_{FC}}L/c$ is the time delay introduced by the FC, L and ${n_{FC}}$ are the length and refractive index of the FC, respectively, and ${\tau _\theta }$ is the time difference for the echo signals arriving at the two antennas.

The reference optical carrier is phase modulated through the upper and lower sub-PMs in the DDMZM, and the 1st-order modulation sidebands are produced. The output of the DDMZM is received by the photodetector (PD2) to realize optical-to-electrical conversion and then filtered by a low-pass filter (LPF). When the noncooperative target is located in the positive direction of the radar sightline, i.e., $0^\circ < \theta \le 90^\circ$, the output after LPF can be described by

(6)$$\begin{aligned}{I_{LPF}} &\propto {J_1}^2(m){J_1}^2({m_U}) \cdot \cos \{{4\pi k({\tau_R} + {\tau_{FC}})t + 4\pi {f_0}({\tau_R} + {\tau_{FC}}) - 2\pi k{{({\tau_R} + {\tau_{FC}})}^2}} \}\\ & + {J_1}^2(m){J_1}^2({m_L}) \cdot \cos \{{4\pi k({\tau_R} + {\tau_\theta })t + 4\pi {f_0}({\tau_R} + {\tau_\theta }) - 2\pi k{{({\tau_R} + {\tau_\theta })}^2}} \}\end{aligned}$$

When the noncooperative target is situated in the negative direction ($- 90^\circ \le \theta \le 0^\circ$), the LPF output signal can be written as

(7)$$\begin{aligned}{I_{LPF}} &\propto {J_1}^2(m){J_1}^2({m_U}) \cdot \cos \left\{ \begin{array}{l} 4\pi k({\tau_R} + {\tau_{FC}} + {\tau_\theta })t + 4\pi {f_0}({\tau_R} + {\tau_{FC}} + {\tau_\theta })\\ - 2\pi k{({\tau_R} + {\tau_{FC}} + {\tau_\theta })^2} \end{array} \right\}\\ & + {J_1}^2(m){J_1}^2({m_L}) \cdot \cos \{{4\pi k{\tau_R}t + 4\pi {f_0}{\tau_R} - 2\pi k{\tau_R}^2} \}\end{aligned}$$

where ${m_U} = \pi {V_{RU}}/{V_{\pi 2}}$ and ${m_L} = \pi {V_{RL}}/{V_{\pi 2}}$ are modulation indexes of the upper and lower sub-PMs, in which ${V_{RU}}$ and ${V_{RL}}$ are the amplitudes of the signals that are applied to the upper and lower sub-PMs, respectively, and ${V_{\pi 2}}$ is the half-wave voltage of the DDMZM.

By dechirp processing of the upper and lower sub-PMs, the low-frequency signals at ${f_1}$ and ${f_2}$ can be obtained after the LPF. When $0^\circ < \theta \le 90^\circ$, these two frequencies can be expressed as

(8)$$\left\{ \begin{array}{l} {f_1} = 2k({\tau_R} + {\tau_{FC}})\\ {f_2} = 2k({\tau_R} + {\tau_\theta }) \end{array} \right.$$

For the case of $- 90^\circ \le \theta \le 0^\circ$, the corresponding frequencies for the output signals are

(9)$$\left\{ \begin{array}{l} {f_1} = 2k({\tau_R} + {\tau_{FC}} + {\tau_\theta })\\ {f_2} = 2k{\tau_R} \end{array} \right.$$

For the positive direction, ${f_2}$ is less than ${f_1}$ when $\theta$ is 0°, and ${f_2}$ will gradually approach ${f_1}$ as the angle increases. Equation (9) shows that ${f_2}$ is always smaller than ${f_1}$ in the case that the target is travelling in the negative direction. In actual applications, a spectrum with two peaks is obtained. To determine which peak is ${f_1}$, it is required that ${f_2}$ cannot exceed ${f_1}$ (i.e., the FC should satisfy ${n_{FC}}L > d$). In addition, since ${f_1}$ and ${f_2}$ will be used for the calculation of the target direction and distance, this qualification also avoids ambiguities in direction and distance measurements. Then, positive or negative $\theta$ can be distinguished by comparing ${f_1} - {f_2}$ and ${f_{FC}}$, where ${f_{FC}} = 2k{\tau _{FC}}$ is the frequency variation introduced by FC. The specific determination guidelines are as follows: when ${f_1} - {f_2} < {f_{FC}}$, $\theta > 0^\circ$; otherwise, $\theta \le 0^\circ$.

Let ${f_\theta }\textrm{ = }2k{\tau _\theta }$ and ${f_R}\textrm{ = }2k{\tau _R}$ be the frequencies associated with the target direction and distance information, respectively. After distinguishing the positive and negative values for $\theta$, for different cases, ${f_\theta }$ and ${f_R}$ can be expressed, respectively, as

(10)$$\left\{ \begin{array}{lc} {f_\theta } = {f_2} - {f_1} + {f_{FC}} , {f_R} = {f_1} - {f_{FC}} & ({0^\circ < \theta \le 90^\circ } )\\ {f_\theta } = {f_1} - {f_2} - {f_{FC}} , {f_R} = {f_2} &({ - 90^\circ \le \theta \le 0^\circ } )\end{array} \right.$$

Then, the direction and distance of the noncooperative target can be deduced by

(11)$$\left\{ \begin{array}{l} \theta = \arcsin \left( {\frac{{c{f_\theta }}}{{2kd}}} \right)\\ R = \frac{c}{2} \cdot {\tau_R}\textrm{ = }\frac{c}{{4k}}{f_R} = \frac{{cT}}{{2B}}{f_R} \end{array} \right.\textrm{ }$$

where $B = 2kT$ is the instantaneous bandwidth of the frequency-doubled detection signal.

The direction and distance can be calculated according to Eq. (11), all of which can be combined to confirm the spatial location of a noncooperative target.

Next, the range resolution and direction resolution of system are analyzed. The theoretical range resolution ${R_{RES}}$ of system can be denoted by [16]

(12)$${R_{RES}} = \frac{{cT}}{{2B}} \cdot \Delta {f_{\min }} = \frac{c}{{2B}}$$

where $\Delta {f_{\min }} = 1/T$ is the spectral resolution.

The theoretical direction resolution is

(13)$${\theta _{RES}} = \arcsin (\frac{c}{{2kd}} \cdot \Delta {f_{\min }}) = arcsin(\frac{c}{{Bd}})$$

As can be seen from Eqs. (12) and (13), an LFM detection signal with large bandwidth can improve both the range resolution and direction resolution. In addition, the direction resolution can also be enhanced by increasing the baseline length.

Meanwhile, as shown by Eq. (11), for a certain frequency detection error, the length of the baseline will impact the direction measurement error. To select a suitable baseline length, the influence of d on the direction measurement error is analyzed by simulation. Here, the frequency measurement error associated with the target direction information is set to 100 kHz, and the baseline length is varied from 0.8 m to 8.0 m. The relationship between the direction measurement error and baseline length is shown in Fig. 3. As the baseline length increases, the directional measurement error gradually decreases. When d >4.0 m, the change in the system measurement error tends to be relatively flat. In addition, an overlong baseline can lead to a serious power imbalance for the RF drive for the upper and lower sub-PMs. Accordingly, the signal-to-noise ratio deteriorates, and high detection accuracy cannot be ensured. Taking the above factors into account, we chose a baseline length of 4.5 m as the experimental condition. To guarantee ambiguity-free measurements, the optical length of the FC needs to be greater than 4.5 m accordingly.

Fig. 3. Curve for the directional measurement error versus baseline length d

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Compared with the conventional phase-based method for direction measurement, the baseline length d for two receive antennas does not need to be less than half the wavelength. Therefore, large aperture antennas can be chosen for greater gain and signal-to-noise ratio. This will largely improve the detection distance and range accuracy of the system.

3. Experiments

To analyze the performance of the proposed simultaneous distance and direction detection method, an S-band radar system was built according to the scheme shown in Fig. 1. The wavelength and optical power of the light source (NKT Photonics, Koheras BASIK) were 1550.12 nm and 14 dBm, respectively. This optical carrier is injected into an MZM (Sumitomo, T-MXH1.5DP-40PD-ADC-LV-S-O) with a half-wave voltage of 3.5 V at 40 GHz. A linear frequency modulation signal is generated by an arbitrary waveform generator (AWG, Tektronix 70002A). Its center frequency, instantaneous bandwidth, and duration are 1.5 GHz, 1 GHz, and 1 μs, respectively. This signal is amplified by a gain-tunable wideband amplifier and applied to the RF port of the MZM. The MZM is biased at the null point to produce ±1-order optical sidebands. The output signal of the MZM is amplified by an EDFA and then split into two beams by an OC. One beam is sent into PD1 (CONQUER Inc, 10 GHz) to realize frequency doubling. The frequency-doubled signal is detected by an electric spectrum analyzer (Keysight, N9020A), and the spectrum is shown in Fig. 4(a). To obtain more characteristics of the signal, a 25 GSa/s digital phosphor oscilloscope (Tektronix, DPO 70604C) was used to capture the signal. The time-frequency characteristics for the signal were obtained by a short-time Fourier transform (STFT), as shown in Fig. 4(b). Its center frequency, instantaneous bandwidth, and duration are 3 GHz, 2 GHz (2-4 GHz), and 1 μs, respectively. The center frequency and bandwidth are doubled, and the signal has excellent linearity. After amplification and filtration by a wideband gain tunable microwave amplifier and an electrical bandpass filter (2-4 GHz@3 dB), the signal is applied onto a wideband transmit antenna to launch into free space as a radar detection signal.

Fig. 4. (a) The spectrum for the transmitted signal; (b) The time-frequency characteristics of the signal.

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A trihedral corner reflector (TCR) is placed at a certain distance in the middle of the radar sightline as the detection target. In the direction detection, the incoming signal commonly demands to be a plane wave. Therefore, the condition of far-field approximation needs to be satisfied. In this case, a time delay equivalently exists between two received signals when the target locates at a certain direction. Here, limited by the laboratory space, direction simulation cables are applied to introduce different time delay [27]. Specifically, the echo signal from the target is collected by a wideband receive antenna and amplified by a low-noise amplifier (LNA, SHF S807C). Then, the amplified signal is divided into two paths via an electrical coupler (EC). One is connected with different lengths of DSC to simulate the echo signal for different angles of the target, and the other is used directly as a receiving driving signal. The positive or negative direction can be simulated by adding the DSC to different ports of the EC, as shown in Fig. 5. It is noted that one of these drive signals is passed through an FC before loading to the upper sub-PM. To balance the drive signal power of the upper and lower PMs in the DDMZM, a tunable electrical amplifier is utilized after FC.

Fig. 5. Experimental diagram for the different target directions. (a) Positive direction; (b) Negative direction. EC: electrical coupler; DSC: direction simulation cable; EA: electrical amplifier.

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At the receiver, the other output signal for the OC is injected into the DDMZM (Fujitsu, FTM7937EZ) as the reference optical signal. The two echo signals after the direction simulation are loaded on the upper and lower sub-PMs of the DDMZM. Since the propagation velocity of the cables is not exactly the same, the lengths of the DSCs were recalibrated. The lengths of the calibrated DSCs are 24.1 cm, 63.9 cm, 101.1 cm, 127.6 cm, 165.4 cm, 203.3 cm, 228.7 cm, 251.8 cm, 277.2 cm and 286.5 cm. Both the upper and lower sub-PMs for the DDMZM are biased at quadrature points by tuning the DC bias voltages. The output signal of the DDMZM is photoelectrically converted by the PD2 (CONQUER Inc, 10 GHz). The dechirp signal of interest was chosen by a low-pass filter (Mini-Circuits, SLP-100+). Then, a 4 GSa/s digital storage oscilloscope (DSO, Agilent Technologies, DSO7104B) was used to sample the output dechirp signal. Finally, a fast Fourier transform (FFT) was performed on the dechirp signal to obtain its spectral information.

The simultaneous distance and direction detection performance for the proposed system was evaluated. First, the situation when the target is located at positive angles is analyzed. The TCR was set at a distance of 110.0 cm from the center of the radar sightline, and DSCs were added between the EC and the lower sub-PM of the DDMZM, as shown in Fig. 5(a). The spectra of the dechirped signals were recorded for different lengths of added DSCs. Figures 6(a) and 6(b) show the spectra for the dechirped signals when the DSCs are 0 cm and 286.5 cm, respectively. The value of ${f_{FC}}$ can be obtained from the difference of the two peaks in Fig. 6(a). Figure 6(c) shows the overall spectrum distribution after inserting different DSCs. Due to the introduction of FC, the value of ${f_1}$ is constantly greater than ${f_2}$. Therefore, we can judge that in each spectrum figure, the peak on the left is ${f_2}$ and the peak on the right is ${f_1}$. Since the target is located at a fixed distance in the positive direction, ${f_1}$ only contains the distance information for the target, while ${f_2}$ is related to both the distance and the direction information for the target. As the angle between the target and the radar sightline increases, the value of ${f_1}$ remains constant, and the value of ${f_2}$ increases continuously. Furthermore, the value of${f_1} - {f_2}$ is always less than ${f_{FC}}$, which can be used in practical applications to adjust the positive or negative direction of the target.

Fig. 6. (a) and (b) Spectra recorded for directional simulation cable lengths of 0 cm and 286.5 cm, respectively; (c) the overall spectrum distribution.

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Combining the spectrum information in Fig. 6(c), the distance and direction of the target can be calculated by using Eq. (11). The experimental measurement distance between the target and the radar antenna is 108.8 cm, which is similar to the actual value of 110.0 cm. Under different DSCs, the measured direction results are shown in Fig. 7. It can be observed that the measured directions have good agreement with the actual direction introduced by the cables, and the system can perform positive direction detection from 0° to 72.5°. The average and maximum measurement errors for the direction are 0.7° and 1.6°, respectively.

Fig. 7. Measured and actual data for targets in the positive direction

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The theoretical range resolution is 7.5 cm under our experimental condition. According to Rayleigh criterion, the range resolution can be calculated through the 3-dB bandwidth of the dechirped signal [28] as shown in Fig. 6(b). The 3-dB bandwidth is 1.68 MHz, corresponding to a range resolution of 12.6 cm. Similarly, the angle resolution can also be obtained by the 3-dB bandwidth of the dechirped signal. The calculation result shows that the angle resolution is 3.2°, which is close to the theoretical value of 1.9°.

Next, the detection capability for the target in the negative direction is evaluated. To simulate this situation, DSCs were added between the EC and the upper sub-PM of the DDMZM, as shown in Fig. 5(b). Here, the TCR is placed at a distance of 80.0 cm from the middle of the radar sightline. The lengths of the DSC remain the same as before. Figures 8(a) and 8(b) denote the spectra for the dechirped signals with direction simulation cables of 0 cm and 286.5 cm, respectively. The spectra are measured and illustrated in Fig. 8(c) for all DSCs used. In this case, ${f_1}$ is determined by the distance and direction information for the target, and ${f_2}$ is only influenced by the distance to the target. As shown in Fig. 8(c), when the target is located at a fixed distance in the negative direction, as the angle of the target grows, ${f_2}$ remains a fixed value, and ${f_1}$ increases continuously. The value of ${f_1} - {f_2}$ is always greater than ${f_{FC}}$, which is quite different from the situation with a positive angle and can be applied to confirm that the angle is negative.

Fig. 8. (a) and (b) Spectra for directional simulation cable lengths of 0 cm and 286.5 cm, respectively; (c) the overall spectrum distribution.

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According to the frequency distribution, the distance and direction information of the noncooperative target can be deduced. The distance between the radar antenna and the target is measured as 82.5 cm, which is quite close to the real value of 80.0 cm. As the DSC changes, the detected results for the target direction are described in Fig. 9. We can see that direction detection from -72.5° to 0° is achieved for the negative direction. The experimental results are consistent with the actual angles introduced by the DSC. The average direction measurement error is 0.6°, and the maximum measurement error is 1.1°.

Fig. 9. Measured and actual data for targets in the negative direction

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In the above experiments, we have verified the detection capability of the proposed method for targets in positive and negative directions, respectively. To make the measurement comprehensive, larger range detection for targets located at 80 cm and 110 cm are carried out. Combining the distance and direction measurement results obtained for the targets, the spatial position localization map for the noncooperative target is plotted in Fig. 10. In the range of -72.5° to 72.5°, the proposed system can be used to simultaneously detect distance and direction. The experimentally measured distance is similar with the actual distance, the average and maximum directional measurement error are 0.6° and 1.6°, respectively. Therefore, the spatial location for the noncooperative target can be accurately located by the proposed method.

Fig. 10. Spatial position localization map of the noncooperative target

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Reaction speed is very important for radar. Once the receiver receives the echo signals, the dechirp process can be realized fast by the microwave photonic link. The frequencies of the dechirped signals are determined by the distance and direction of target. The subsequent treatment is mainly composed of ADC, single digital Fourier transform (DFT), signal extraction etc. Among them, DFT is the most time consuming, Digital Signal Processing (DSP) is one of the best tools to improve the detection speed.

4. Conclusion

A simultaneous detection method of distance and direction for the noncooperative target based on microwave photonic radar is proposed and established. At the transmitter, a frequency-doubled LFM signal is produced through a MZM, and it is emitted as the detection signal. The optical reference signal is also generated for the receiver. A fixed-length cable is specifically inserted to determine the positive or negative direction of the noncooperative target. The two frequencies associated with multiple information of target are obtained by dechirp processing of the echo signal. Then, the positive or negative angles can be distinguished by comparison of the frequency difference for the two dechirp signals and the frequency variation introduced by the FC. After that, both the direction and distance information for the target can be extracted. The proposed approach possesses several attractive features. Primarily, the spatial location for the noncooperative target can be confirmed by the function integration of the simultaneous distance and direction measurement in a compact microwave photonics link. The dechirp process and a FC are combined to decouple the distance and direction information. In comparison to the links based on single function, more information of the target can be supplied. Secondly, ambiguity of positive and negative direction is solved by using the FC, and detection of large angles for space target has been verified. Furthermore, comparing with the direction detection method based on phase difference, our frequency-based measurement is almost insensitive to the power fluctuation of the laser or the bias drift of the modulator; meanwhile, the baseline length can be chosen more flexibly without the limitation of less than half the wavelength of the echo signal. Therefore, large antennas can be chosen to improve the detection distance and range accuracy. The proposed approach can supply integrated and large-range spatial location scheme for the detection or tracking of noncooperative target.

Funding

Beijing Municipal Natural Science Foundation (4202001); National Natural Science Foundation of China (61771438, 61871007).

Acknowledgments

The authors thank Prof. Pingxue Li, Dr. Chuanfei Yao, Prof. Yanrong Song and Dr. Jinrong Tian for their assistances on experiments.

Disclosures

The authors declare no conflicts of interest.

Data availability

Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.

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Simultaneous detection of the distance and direction for a noncooperative target based on the microwave photonic radar

Abstract

1. Introduction

2. Principle

3. Experiments

4. Conclusion

Funding

Acknowledgments

Disclosures

Data availability

References

Data availability

Cited By

Figures (10)

Equations (13)

Optics Express