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Abstract
Radiofrequency (RF) channelization has potential high frequency and wideband advantages in frequency-domain channel segmentation and down-conversion reception. In this paper, we propose a compact dual-channel channelizer that can process high-frequency wideband signals. It uses double-polarization double-sideband electro-optic modulation and Hartley structure photoelectric conversion to realize down-conversion channelization of the high-frequency wideband signal. The power matching between two polarization signals can be realized by controlling the modulator bias, so the crosstalk between the two output signals can be suppressed. The proposed channelizer has a compact structure since the electro-optic modulation is based on one single laser and one single integrated modulator. No filters are used in the structure, contributing to a very wide RF operation bandwidth and low constraints of laser wavelength. In the experiment, the single frequency signal pairs from 9 GHz to 15 GHz can achieve an inter-channel image rejection ratio of 53 dB. Furthermore, the channelizer slices multi-octave bandwidth quadrature phase shift keying (QPSK) signals up to 16 GHz with the wideband isolation higher than 10 dB and outputs them to two channels in parallel. The error vector magnitudes (EVM) of 9–17 GHz and 18–26 GHz band QPSK signals are guaranteed to be under 23.58% after channelized separation. To the best of our knowledge, the proposed channelizer provides high inter-channel interference suppression at dual-band adjacent signals with 8 GHz bandwidth for the first time. Therefore, the proposed channelizer has great application value for the reception and processing of millimeter signals in the future.
© 2022 Optica Publishing Group under the terms of the Optica Open Access Publishing Agreement
1. Introduction
In recent years, 5G, beyond 5G, ultrahigh-capacity satellite communication, high-frequency agile radar in electronic warfare, and many other industries demand the signals to develop towards higher-frequency, larger-bandwidth, multi-band [1–3]. As a result, there is an urgent need for high-performance receivers with large instantaneous bandwidth, high resolution, large dynamic range, and the ability to process multi-frequency signals. Channelization is an effective way to meet this demand by slicing the received wideband RF signal into multiple narrow sub-signals, which not only reduces the sampling rate requirement of the analog-to-digital converter (ADC) but also relieves the digital signal processing (DSP) complexity. The traditional analog channelized receiver is composed of microwave devices, which leads to extremely high losses in high-frequency bands and cannot achieve uniform high performance over one or several octaves. These analog channelized receivers are gradually replaced by the digital channelized receiver [4]. Digital channelization, based on the design of digital filter arrays, can divide channels and switch outputs flexibly by appropriately processing integrated chips such as FPGA [5]. However, using DSP to mine target information with high fidelity poses great challenges for high-speed processing chips, high-rate sampling modules, and high-storage memory.
Microwave photonics combines the advantages of broadband, low-loss, flat spectrum response optical waves, and narrowband fine control of microwaves. It overcomes the electronic bottleneck of traditional broadband analog or digital processing and provides a new solution for radio frequency signal generation, transmission, and processing [6,7].
Therefore, the channelization based on microwave photonics has been widely concerned by scholars, and several methods have been reported [8–17]. One of these methods is based on a group of optical filters with consecutive passbands, such as Bragg grating Fabry-Perot cavity, forming an optical filter array that slices a broadband signal into lots of sub-signals [8–10]. Such a scheme only needs a single optical carrier. However, optical filters with flat-top and steep-edge frequency responses are typically required, which makes device fabrication considerably difficult. Another method multicasts a broadband RF signal over an optical frequency comb (OFC) and then uses a comb filter with a free spectral range slightly detuned with comb line spacing to generate narrowband signals [11]. The third method uses dual coherent OFCs, one of which is used to form a series of copies of the broadband signal, and the other to label the cutting position for the broadband signal, avoiding conventional fine spectral filtering [12–14]. The optical combs are separated by a wavelength division demultiplexer (WDM-DEMUX) to realize channel division after photoelectric detection. Restricted by the flatness and number of the optical comb lines, the acceptable RF range is limited. Coherent detection between the chirp pulse and its delay can generate a tunable local oscillator (LO) up to tens of GHz, allowing channelization in different frequency bands. However, the frequency of the LO needs to meet an integer multiple of the pulse repetition frequency, which undoubtedly increases the complexity of the system [15]. A pair of dispersion-compensating fibers (DCFs) with complementary dispersion coefficients are used to stretch and compress the optical pulse for frequency-time mapping. So, by mapping the frequency information of the broadband signal to the time-domain information of the optical carrier, spectral segmentation can be realized by the time-domain sampling of Fourier-transformed pulses. However, the system performance can be affected by the non-uniformity of the Bragg grating intensity response [16,17]. The above methods use complex structures to complete multi-channel channelization and are not conducive to photonic integration. At present, the state-of-art commercial FPGA digital ADCs have a sample rate of up to 10 GSa/s and can process signals with a bandwidth of 5 GHz, which will have higher parameters in the future. The current bandwidth demand for practical applications in the millimeter band is not too wide, more than ten GHz is already abundant. Consequently, based on the existing technological level and application requirements, the dual-channel channelizers are already practical and should be studied most.
The dual-channel channelizer can be realized by the extension of image rejection technology. Various image rejection methods have been proposed such as cascaded phase modulator (PM) and Mach-Zehnder modulator (MZM), asymmetric Mach-Zehnder interferometer (AMZI), etc [18,19]. These methods rely on optical filters or WDM-DEMUX to filter out a sideband or separate the upper and lower sidebands. The bandwidth is limited, and the cascading method produces unnecessary harmonic interference [18,19]. Making use of a Sagnac loop and an I/Q balanced detector, the image interference can be suppressed by rotating the polarization controller. However, the image rejection ratio (IRR), a characterization of the crosstalk isolation of the channelizer, is susceptible to amplitude and phase imbalance [20,21]. Therefore, to meet the current application requirements and make up for the shortcomings of the above methods, we focus on the research on the concise structure of the dual-output channelizer.
In this paper, a broadband dual-channel channelizer based on an integrated dual-polarization dual-parallel Mach-Zehnder modulator (DP-DPMZM) is presented and verified. The proposed system has three merits: first, the system is very simple and compact since it is based on an integrated modulator and the orthogonal branches are constructed with two polarization. Second, it introduces a pair of orthogonal LO signals, eliminating the need for sharp roll-off edge filters, and the high-frequency and large-bandwidth signals can be orthogonally down-converted. The down-converted signals within a wide frequency range can be channelized through a dual-output 90° hybrid coupler (HC). Third, the branch power imbalance is reduced by controlling the parallel link power, which is achieved by adjusting the phase difference between the sub-MZMs under the two polarization states of the modulator. We conducted an experimental verification. The single frequency signal pairs with an operating bandwidth from 9 GHz to 15 GHz are successfully channelized, showing channel isolation with an IRR up to 53 dB. Furthermore, we carried out dual-channel channelization of dual-band QPSK signals with the bandwidth of 8 GHz with center frequencies in the K and Ku band, the crosstalk suppression between channels is guaranteed to be above 10 dB, and the error vector magnitudes (EVM) after demodulation are under 23.58%.
2. Principle
Figure 1(a) shows the schematic diagram of the proposed dual-channel channelizer. An optical carrier (OC) from an LD is fed to the DP-DPMZM via a polarization maintaining fiber. The angular frequency of the optical carrier is ${\omega _c}$. The DP-DPMZM, whose detailed structure is shown in Fig. 1(b), is a commercially available integrated modulator, which consists of two dual-parallel Mach-Zehnder modulators (DPMZMs) including two push-pull MZMs, a 90° polarization rotator (PR), and a polarization beam combiner (PBC). We feed a dual-band wideband signal to verify the performance of the proposed dual-channel channelizer. The two center frequencies of the wideband signal are symmetrical about the frequency of the LO signal, denoted as ${\omega _{s,lef}}$ and ${\omega _{s,rig}}$. Given that the equivalent baseband signal waveforms are ${s_{lef}}(t )$ and ${s_{rig}}(t )$, respectively. ${V_{LO}}$ and ${\omega _{LO}}$ are the amplitude and angular frequency of the LO signal. Then the wideband signal is injected into the upper arm MZMs (MZM1 and MZM3). The LO signals are respectively applied to the lower arm MZMs (MZM2 and MZM4) through the 90° HC, avoiding the direct electrical tuning of the broadband input signal. The modulated optical signals of each MZMs can be expressed as the following
Fig. 1. (a) The schematic diagram of the proposed dual-channel channelizer; (b) The detailed structure of the DP-DPMZM; LD: laser diode; DP-DPMZM: dual-polarization dual-parallel Mach-Zehnder modulator; PBC: polarization beam combiner; PR: polarization rotator; PC: polarization controller; PBS: polarization beam splitter; PD: photodetector.
By adjusting the bias voltages of the MZMs to achieve ${\varphi _1} = {\varphi _2} = {\varphi _3} = {\varphi _4} = \pi$, we implement the carrier-suppressed dual-sideband (CS-DSB) modulation. Based on the Jacobi-Anger expansion and considering small-signal modulation only, Eq. (2) can be expanded as
Then the modulated optical signal is sent to a PC. By adjusting the PC, the modulated signal is divided into X-Pol and Y-Pol of the PBS. Under the small signal modulation, ${J_1}({{m_{lef}}{s_{lef}}(t )} )\approx {J_1}({{m_{lef}}} )\cdot {s_{lef}}(t )$ and ${J_1}({{m_{rig}}{s_{rig}}(t )} )\approx {J_1}({{m_{rig}}} )\cdot {s_{rig}}(t )$, ignoring the DC components and the third-order distortions, the photocurrents generated by the PDs can be described by
From Eqs. (4) and (5), we can draw two conclusions. First, the phase shifts caused by the main bias voltages of the DPMZMs in both X and Y polarization are transferred to the amplitude of the output signals. Second, the broadband signals located to the left and right sides of the optical LO after photoelectric detection are simultaneously down-converted to the same mid-frequency band, resulting in aliasing. To avoid overlapping, the two signals are combined using an electrical low-frequency 90° HC, and the final two output signals can be expressed as
It is worth stating that in a practical experiment system, the amplitude, phase, and delay imbalance are important factors affecting channel interference. In the experiment, several factors cause the amplitude imbalance, such as the different modulation efficiency under two branches of the electro-optical modulator. The power imbalance can be reduced by properly adjusting the main bias voltage, which we will show in the following section. The phase imbalance mainly originates from the imprecise phase shift of the 90° HC. One of the 90° HC is used to introduce a phase shift to the LO signal, changing the phase of a single frequency point and avoiding tuning of wideband signals. Another one is used to change the phase of the narrow-band low-frequency signal, which greatly reduces the device requirements. The two optical lengths may be different, leading to optical delay. Such delay will introduce phase difference and reduce system wideband performance. We use a dual-channel optical delay line in the experiment to compensate for the phase imbalance and the absolute delay.
3. Experiments and results
A proof-of-concept experiment is implemented according to Fig. 2. An optical carrier with a wavelength of 1550 nm and the average power of 13 dBm emitting from a tunable laser source is fed to a DP-DPMZM (Fujitsu FTM7977HQA). The linewidth of the LD is about 100 kHz. The LO signal generated by an analog signal generator (ASG, Anritsu 68047C), is split into two parts through a 90° HC and injected into the MZM2 and MZM4. Wideband RF signals with the peak-to-peak voltages of 400 mV generated by a 120 GSa/s arbitrary waveform generator (AWG, Keysight M8194A) drive the MZM1 and MZM3. The PC and the PBS are employed to demultiplex the polarized signals and then feed the polarization demultiplexed signals to two PDs with 30 GHz bandwidth (Optilab PD-30) for square-law detection. The optical variable delay lines are used to compensate for the absolute delay between the parallel links. Finally, the signals are output through two ports of a low-frequency 90° HC. In addition, an oscilloscope (Agilent DSO-X 93204A) with a sampling rate of 80 GSa/s digitalizes the output signal for future offline processing.
Fig. 2. Experimental setup of the proposed dual-channel channelizer. AWG: arbitrary waveform generator; ASG: analog signal generator; EDFA: erbium-doped fiber amplifier; OVDL: optical variable delay line.
First, we investigate the performance of the orthogonal links with power controllable capability. The other ASG (Keysight N5173B) replaces the AWG to generate the single frequency signal. The LO signal is set at 12 GHz with a power of 6 dBm. When a single frequency RF signal with a frequency of 9.5 GHz and the power of 10 dBm is fed, Figs. 3(a) and 3(b) show the optical spectra of the generated signal before and after its depolarization. We can see that the power of the optical carrier is nearly 30.79 dB lower than the modulated 1st -order sidebands, indicating that the CS-DSB is achieved. Figure 3(b) shows the optical spectra of the X and Y polarization after the PBS. The signal sideband has a nearly 0.4 dB power difference, which is determined by the modulation efficiency of DP-DPMZM. Subsequently, we adjust the main bias voltage applied to DPMZM-x and DPMZM-y to observe and record the power of the IF signals after two PDs, as shown in Fig. 4. The blue line represents the x-pol branch signal power-vary ${V_x}$ and the red line represents the y-pol branch signal power-varying ${V_y}$. We can see that the output sideband power of the branches is controllable by the bias voltages, that is, the link amplitude imbalance can be solved by fine-adjusting the voltage. So the 0.4 dB power difference can be mitigated. With such little difference, the optical power of two polarizations can be regarded as the same. It shows that the two polarizations are well separated.
Fig. 3. (a) The optical spectra of the signal after EDFA and (b) X and Y polarized optical spectra after the PBS.
Fig. 4. Measured the 2.5 GHz signal power obtained from the branch by adjusting the main voltage in x-polarization and the branch by adjusting the main voltage in y-polarization.
Then, the isolation of the proposed dual-channel channelizer is verified under the single frequency test. The LO signal frequency is set to be 12 GHz, and the 9.5 GHz RF1 signal is fed. Figure 5(a) shows that the IF signal power after two PDs is −22.75 dBm and −22.99 dBm, respectively. It should be noted that the output power difference after two PDs is within 1dB by adjusting the bias voltages of the modulator. After the electrical quadrature hybrid, the signal power output by channel 1 becomes −66.55 dBm. At the same time, the power of the signal in channel 2 is −16.63 dBm, increasing nearly 6 dB gain compared to a single link with about −22 dBm signal power. So, we can see that the RF1 signal is suppressed on channel 1 and enhanced on channel 2.
Fig. 5. Electrical spectra of output signals of PDs and combined IF signals after ${90^ \circ }$HC. (a): PD outputs and channel 1, channel 2 IF signal with RF signal set at 9.5 GHz. (b): PD outputs and channel 1, channel 2 IF signal with RF signal set at 14.5 GHz.
Similarly, when feeding into an RF2 signal with a frequency of 14.5 GHz, as we can see in Fig. 5(b), the signal powers of the two branches are −22.01 dBm and −22.79 dBm, respectively. The final output signal power of channel 1 increases to −17.53 dBm, while the signal output of channel 2 decreases to −65.79 dBm, indicating that the RF2 signal is enhanced on channel 1 and suppressed on channel 2. In total, in channel 1, the signal at 9.5 GHz is suppressed and the signal at 14.5 GHz is enhanced, while in channel 2 the condition is the contrary. The inter-channel IRR of the proposed dual-channel channelizer can reach 49.92 dB.
To demonstrate the inter-channel IRR performance at different frequencies, the RF signal frequency is tuned from 9 to 15 GHz with a step of 500 MHz and the LO is 12 GHz. The electrical spectra of the IF signal are shown in Fig. 6. When the RF signal goes from 9 GHz to 11.5 GHz, the output signal is suppressed below −74 dBm. When the frequency of the RF signal is from 12.5 GHz to 15 GHz, the IF signal power changes to −21 dBm, demonstrating that the inter-channel IRR can reach as high as 53 dB.
Fig. 6. Electrical spectra of IF signals from Channel 1 within the frequency from 0.5–3 GHz when the LO is 12 GHz.
The SFDR of the proposed dual-channelizer is measured when channel 1 is enabled. The two-tone RF signal with frequencies of 9.5 GHz and 9.51 GHz is generated by the ASGs. The LO is fixed at 12 GHz and generated by the vector network analyzer (VNA). The output power is set at −5 dBm and the electrical amplifier of 18 dB gain is used. When the signals are applied to the channelizer, the fundamental to IMD3 power ratio (FIPR) and SFDR are tested respectively. The resolution bandwidth (RBW) of the electrical spectrum analyzer (ESA) is 20 kHz. Figures 7(a), and (b) show the measured FIPR and SFDR. As can be seen that the FIPR is 48.5 dB, and the SFDR of the channelizer is calculated to be 100.01 $\textrm{dB} \cdot \textrm{H}{\textrm{z}^{2/3}}$ measuring the noise floor at −143.48 dBm/Hz.
To better verify the spectra-cutting ability of the broadband signal, we carried out a comparative experiment. The QPSK signal generated by an AWG has a symbol rate of 4 GBaud. The pulse shaping filter at the transmitter and the matched filter at the receiver are the same root-raised-cosine filter with a roll-off factor of 0.01. A sequence with a total length of 384 is placed at the front of the QPSK signals. It consists of 6 repeated const amplitude zero auto-correlation (CAZAC) sequences. The training sequence is used for channel frequency response calculation and inter-channel response interference estimations. The QPSK signals are then used to evaluate the performance of our system. After equalization, the EVM of the QPSK signal is calculated from 400000 QPSK symbols for a more representative result. Figures 8(a–c) shows the electric spectra of different wideband signals with optimal input power. The frequency of the LO signal is 21.5 GHz. Only the wideband signal with a center frequency of 19 GHz is injected, signal spectra from two channels after 90° HC are shown in Fig. 8(d). It can be seen that the power of the signal is about −28 dBm in channel 2, while in channel 1, the power of the signal is lower than −41 dBm. It means that the signal on the left side of the LO signal is down-converted and stored in channel 2. Due to the optical amplification, the noise is raised to −55 dBm, affecting the system performance. The constellation of vector signals from channel 2 is shown in Fig. 8(i). The calculated EVM is 11.01%. Only the wideband signal centered at 24 GHz is applied, the spectral of two outputs are shown in Fig. 8(e). The output signal power of channel 1 is 13 dB higher than that of channel 2, indicating that the signal located on the right side of the LO is down-converted and stored in channel 1. The constellation of the vector signal is shown in Fig. 8(j), the EVM is 13.50%. When a dual-band signal is fed, the broadband signals complete spectrum cutting and output in both channels in parallel. And the corresponding constellation diagrams are shown in Figs. 8(k) and 8(l). The EVMs calculated is 17.04%, and 18.25%, respectively. Then, we measure the intra- and inter-channel interference responses in Fig. 8(f). It can be seen that the intra-channel (channel 1 to channel 1 and channel 2 to channel 2) interference coefficient curves are quasi-rectangular functions. That is, due to the uneven phase and amplitude in orthogonal branches and the system nonlinearity, the curves appear the peaks. The inter-channel (channel 1 to channel 2 and channel 2 to channel 1) interference coefficient curves are close to zero, indicating that the proposed dual-channel channelizer can work with good performance in a large bandwidth.
Fig. 8. Measured electrical spectra of the input signal (a)-(c); The electrical spectra of the output signal from channel 1(blue line) and channel 2 (red line) when only feeding a wideband RF signal with 4 GHz bandwidth in the left-band (d), right-band (e), respectively; Measured the intra- and inter-channel interference responses (f); The constellation diagrams of IF signals from the channel 2 when feeding left-band signal (i), the channel 1 when feeding right-band signal (j) and the two channels (k)-(l) when feeding dual-band signals.
Then, to evaluate the broadband dual-channel interference suppression capability, we first send a wideband QPSK signal at 9–17 GHz into the RF port, then replace it with a signal at 18–26 GHz, and at last, we send the combination of two signals. The electrical spectra are shown in Figs. 9(a–c). The LO is fixed at 17.5 GHz. When the QPSK signal with a frequency from 9 GHz to 17 GHz is injected, the two-channel outputs are shown in Fig. 9(d). Only the QPSK signal at 18–26 GHz is applied, the two outputs are shown in Fig. 9(e). We can see that the broadband signals on the left and right sides of the LO are output on channels 2 and 1, respectively. And the channel interference suppression is about 10 dB. The constellation diagram of the vector signal is shown in Figs. 9(i–j). The calculated EVMs are 13.13% and 15.32%. When feeding the combined dual-band signal, the channel interference curves are shown in Fig. 9(f), the curves have peak jitter. This is mainly caused by the crosstalk when feeding multi-octave band signals. There are two sources of crosstalk, one originates from signals self-beating, and the other is signals cross-beating. The constellation diagrams of the two-channel QPSK signal are shown in Figs. 9(k) and 9(l), and the EVMs of the signals are 23.58% and 22.68%.
Fig. 9. Measured electrical spectra of the input signal (a)-(c); The electrical spectra of the output signals from channel 1(blue line) and channel 2 (red line) when only feeding a wideband RF signal with 8 GHz bandwidth in the left-band (d), right-band (e), respectively; Measured the intra- and inter-channel interference responses (f); The constellation diagrams of IF signals from the channel 2 when feeding left-band signal (i), the channel 1 when feeding right-band signal (j) and the two channels (k)-(l) when feeding dual-band signals.
4. Conclusion
In conclusion, we propose and demonstrate a photonics dual-channel channelizer based on the image rejection down-conversion technique. The structure not only meets the current bandwidth application requirements but also can be easily integrated since only one commercial modulator is used contributing to a compact system. In addition, the orthogonal branches’ power imbalance can be reduced by adjusting the bias voltages, so that the inter-channel image interference is well suppressed. Moreover, the operating frequency range is wide because no filters are used. A proof-of-concept system has been demonstrated to verify the performance, in which single frequency signal pairs from 9 GHz to 15 GHz can achieve 53 dB inter-channel IRR, and the dual-band QPSK signals with bandwidths up to 8 GHz centered at K and Ku band completely channelized. The EVMs of the QPSK signal after channelization are under 23.58%. The spurious free dynamic range (SFDR) of the proposed system is 100.01 $\textrm{dB} \cdot \textrm{H}{\textrm{z}^{2/3}}$. The proposed photonic dual-channel channelizer has potential applications in future broadband communication, contributing to the integration of multi-band, multi-frequency, and multifunction architecture.
Funding
National Natural Science Foundation of China (62125103, 61821001, 62171050, 62171059).
Disclosures
The authors declare no conflicts of interest.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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