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Abstract
In this work, an ultra-wideband flexible radar absorber with low infrared emissivity for a radar-infrared bi-stealth application utilizing multilayer patterned graphene is proposed. The proposed absorber consists of three layers of graphene films with different patterns, flexible substrates, lightweight foam, and a ground layer. The flexible graphene films, rather than the conventional lumped resistors, are adopted as omnidirectional resistors to achieve dual polarization and flexibility. On the top of the absorber, an infrared shielding layer (IRSL) consists of patterned Indium tin oxide (ITO) separated by a thin foam layer. Due to the low-pass characteristics and the high filling ratio of the top ITO layer, the infrared emissivity of the whole structure is reduced effectively while the radar absorption property is slightly affected. As a result, the 90% absorption band is from 1.96 GHz to 20.72 GHz (fractional bandwidth 165.4%), with a low infrared emissivity of about 0.35. Besides, a miniaturized unit is achieved with the period of 0.079 λl at the lowest absorption frequency, and the oblique angle incidence response is up to 45° for TE mode and 60° for TM mode. A plane and a bending prototype are fabricated and measured, respectively. The screen-printing technology is adopted to print the graphene resistive films, and the measurement results agree well with the simulation.
© 2023 Optica Publishing Group under the terms of the Optica Open Access Publishing Agreement
1. Introduction
With the development of multi-spectrum composite detection technology, the stealth technology of a single frequency band can no longer meet the military needs of current weapons and equipment. Infrared (IR) and radar are the two most commonly used detection methods, for this reason, the task of developing radar/infrared bi-stealth materials is increasingly urgent [1–8]. Radar stealth is achieved by reducing reflected waves and radar cross sections [9–12], and the key to infrared stealth and camouflage is changing the infrared radiation so that it is close to the background or the camouflage target [13–15]. Therefore, radar stealth requires low reflectivity and high absorption, while infrared stealth requires high reflectivity and low absorption (emissivity). However, it is difficult for traditional materials to express high radar absorption and low infrared emission at the same time.
In recent years, metamaterials have attracted great attention in stealth technology [16–18], and THz modulators [19]. By designing specific unit structures artificially, perfect absorption [20], low scattering [21], electromagnetic stealth [22], etc. could be achieved. Graphene exhibits high electron mobility, flexibility and adjustable resistance in the microwave band, which is widely used in absorbers [23–25]. Besides, ITO as a low IR emissivity material with flexibility, transparency and lightweight, has a broad application prospect in the field of infrared and radar stealth [26,27]. At present, the main approach to achieving radar-infrared bi-stealth stealth is to superimpose infrared stealth material and radar absorption material. The key to this method is the infrared stealth layer on the top surface, where not only its own low infrared emissivity needs to be ensured but the maximum transmission of radar waves [28,29]. In this way, the radar waves can be effectively lost in the absorption layer. Besides, the radar absorption and infrared stealth performance can be individually designed and manipulated due to the respective operating methods and working bands. Some metamaterial-based radar-infrared bi-stealth structures have been presented in the open literature to realize wide absorption band [30,31], ultralight [32], flexibility [33–35], optical-transparent [34,36], wide temperature ranges [37] and signature control based on magnetic resonance [38]. In Ref. [34], flexible polyethylene terephthalate (PET) and ITO are used to design a flexible bi-stealth metasurface. However, due to the single loss layer and the curvature limitation of PET, the fractional bandwidth of 90% absorption is only 114.5%. By a multi-layer design, the bi-stealth metasurface proposed in Ref. [31] realized an ultra-wideband microwave absorption from 2.7 GHz to 26 GHz (fractional bandwidth of 162.3%) and a low emissivity of 0.2, while the structure lack of flexibility due to the all-metallic IR layer. Therefore, there is no doubt that ultra-wideband radar-infrared bi-stealth absorber with flexibility characteristic is of great importance to improve the integration with electromagnetic equipment.
In this work, we propose a multilayer radar-infrared bi-stealth structure, which exhibits ultra-wideband microwave absorption covering S, C, X, Ku and parts of K bands, low infrared emissivity, conformal simplicity as well as good incident angle stability. The flexible patterned graphene films are adopted as omnidirectional resistors to achieve ultra-wideband absorption from 1.96 GHz to 20.72 GHz. A high filling ratio ITO IRSL is set on the top of the absorber and the emissivity in the infrared light of 8-14 µm is about 0.35. Due to the miniaturized unit design, the oblique angle incidence response is up to 45° for TE mode and 60° for TM mode, respectively. Besides, the proposed radar-infrared bi-stealth structure is easy to be conformed to the detected surface due to the flexible graphene lossy films rather than the traditional lump resistors. Both simulation and measurement results demonstrate that the proposed radar-infrared bi-stealth structure expresses broad prospects in multispectral stealth applications.
2. Design and analysis
2.1 Geometrical structure and simulation results
The unit cell of the proposed radar-infrared bi-stealth structure is shown in Fig. 1. Three layers of patterned graphene, flexible substrates, lightweight foam and a metal ground compose the ultra-wideband radar absorber. An ITO infrared stealth layer is covered over the absorber and separated by a thin layer of foam. The flexible lightweight Polyvinyl chloride (PVC) (εr1 = 2.7, tan δ = 0.007) is adopted as dielectric layers and three layers of lightweight flexible foam (εr2 = 1.1) are adopted to provide support and coupling between the adjacent layers. Due to the adjustable sheet resistance and the low printing cost, the patterned graphene films are used as the main lossy component with a sheet resistance of 25, 30 and 20 Ω/sq from top to bottom. Compared with the traditional lumped resistor, the graphene film could achieve dual-polarization properties with only one print. The conductive graphene ink was printed on a PET substrate (εr3 = 3.5) with a thickness of 0.075 mm, which is thin enough and makes a negligible effect on the absorption performance compared with the operating wavelength. The physical dimensions of the unit are as follows: h0 = 1 mm, h1= 1 mm, h2= 4 mm, h3= 1 mm, h4= 4 mm, h5= 1.5 mm, h6= 4 mm, a1 = 8.47 mm, a2 = 9.9 mm, a3 = 0.86 mm, l1 = 3.3 mm, l2 = 2.75 mm, w1 = 0.44 mm, w2= 0.55 mm, w3= 0.27 mm, w4= 0.55 mm, w5= 1.32 mm, w6= 0.88 mm, g1 = 0.11 mm, g2 = 0.14 mm, p = 12.1 mm.
Fig. 1. The geometrical structure of the proposed absorber. (a) Schematic of the flexible array. (b) Side view of the unit cell. (c) Top lossy layer. (d) Middle lossy layer. (e) Bottom lossy layer. (f) Infrared shielding layer.
The simulation reflection coefficients under TE and TM normal incident waves are shown in Fig. 2(a). It can be seen that since the patterns of each layer are four-fold symmetric, the reflection coefficient of the proposed absorber is insensitive to the polarization angle of the incident wave. Figures 2(b) and (c) show the simulated reflection coefficient under TE and TM oblique incidence waves at different angles. The oblique incident response of the reflection coefficient remains stable up to 45° for the TE mode and 60° for the TM mode. Besides, a 25 × 25 plane array is simulated in CST Studio Suite under 45° incidence TE waves. The bistatic RCS compared with a perfect electric conductor (PEC) of the same area at different frequencies are as shown in Figs. 2 (d)-(f). Due to the miniaturized unit design of the meander line pattern [25], the RCS reduction compared with PEC is more than 10 dB and a low sidelobe characteristic is realized at frequencies less than 12 GHz as shown in Figs. 2 (d) and (e). With the increase of frequency, the sidelobe gradually becomes higher due to the decreasing operating wavelengths compared with the unit period. Therefore, an obvious sidelobe appears at 16 GHz (λ) with the unit period of 0.65 λ as shown in Fig. 2 (f).
Fig. 2. Simulated reflection coefficient and bistatic RCS of the proposed plane structure. (a) S11 under normal incidence. (b) S11under oblique incidence in TE mode. (c) S11under oblique incidence in TM mode. (d) Bistatic RCS of the plane array and PEC in TE mode under 45° incidence angle at 8 GHz, (e) 12 GHz, (f) 16 GHz.
Furthermore, due to all the materials adopted in the proposed structure such as ITO, graphene film, PET, PVC and foam being flexible, it is easy to be conformed to the detected surface. As shown in Fig. 3(a), an 8 × 10 array conformed to a cylinder with different bending radius (r) is simulated under normal incidence plane waves. The bistatic RCS compared with a PEC is simulated to demonstrate the far-field scattering properties under bending condition, as shown in Figs. 3(b)-(e). It shows that a great RCS reduction could be realized under different bending radius. At 6 GHz as shown in Figs. 3(b) and (d), due to the long operating wavelength, fewer side lobes occur at other oblique received angles. An effective reduction of bistatic RCS is expressed which is larger than 10 dB from -90° to 90°. As shown in Figs. 3(c) and (e), due to the reduced operating wavelength at 16.4 GHz, the electrical size of the conformal array increases and more side lobes appear at the oblique received angles. Besides, with the increase of bending radius and frequency, the RCS reduction performance shows a little deterioration. Even so, the proposed structure keeps a stable RCS reduction close to 10 dB.
Fig. 3. Simulated bistatic RCS of the flexible array and PEC under normal incidence with different bending radius. (a) Simulated array with 8 × 10 unit cells. (b) 6 GHz, r = 70 mm. (c) 16.4 GHz, r = 70 mm. (d) 6 GHz, r = 100 mm. (e) 16.4 GHz, r = 100 mm.
2.2 Infrared shielding layer with low emissivity
To avoid detection by infrared devices, the ideal infrared stealth material should perform the following two characteristics: the low infrared emissivity of the stealth layer itself and the high reflectivity of infrared waves coming from the inner stealth structure. Metal is the ideal infrared stealth material while it is heavy and hard to conformal. For this reason, an ITO film based on PET with a low sheet resistance of 6 Ω/sq is adopted to design the flexible IRSL, which shows similar IR shielding properties to metal. As shown in Fig. 4(a), the IRSL unit is composed of a square ITO patch with a side of a3. Due to the small period compared with the bottom absorber unit, a strong capacitive effect is introduced. Therefore, the IRSL can be equivalent to a spatial low-pass filter in the microwave band as shown in Fig. 4(b). The passband with the insertion loss (IL) less than 1 dB is from 2 GHz to 22 GHz, which covers the absorption band of the bottom absorber. Therefore, the radar waves in the absorption band could transmit the IRSL with an ignorable IL and be lost by the bottom absorber.
Fig. 4. (a) Geometry structure of the IRSL unit. (b) Simulated S21 of the IRSL and S11 with and without IRSL of the whole bi-stealth structure.
The surface electric field distributions of the IRSL at different resonance frequencies are shown in Figs. 5(a) and (b). It can be seen that at 8.2 GHz, the capacitive effect between the adjacent patches is weak and the low frequency makes the surface impedance approaches infinity. Therefore, electromagnetic waves at low frequencies can transmit the IRSL with a slight insertion loss. While the electric field at the adjacent gap is enhanced obviously at 19.4 GHz. The strong capacitive effect and the high work frequency introduce a large insertion loss close to 1 dB. Corresponding to the simulated S11 of the whole bi-stealth structure with and without IRSL, as shown in Fig. 4(b), the two curves gradually deviate as the frequency increases. Therefore, a high microwave transmittance of the IRSL especially at high frequencies needs a weaker capacitive effect and smaller filling ratio of ITO, which means a small a3 and is unfavorable to reducing infrared emissivity. The emissivity of the IRSL ${\varepsilon _1}$ could be calculated by the filling ratio of ITO [39,40]:
where ${\varepsilon _m}$, ${\varepsilon _s}$ represents the emissivity of ITO and PET respectively, and f is the filling ratio of ITO. Generally, ${\varepsilon _m}$ is about 0.05 and ${\varepsilon _s}$ close to 0.9 [41]. Therefore, the larger the filling ratio of ITO, the lower the infrared emissivity which is contrary to radar absorption at high frequencies. According to formula (1), the calculated infrared emissivity is 0.27 with the filling ratio f of 74%.Fig. 5. Surface electric field distribution of IRSL at low and high resonant frequencies. (a) 8.2 GHz. (b) 19.4 GHz.
2.3 Energy loss mechanism and the equivalent circuit model
To visually represent the working mechanism of the proposed absorber, Fig. 6 shows the surface current distribution of each lossy layer at four resonance frequencies. As shown in Figs. 6(a) and (b), in the direction parallel to the incident electric field, the surface current on the top three layers is opposite to that on the metal ground at 2.6 GHz and 8.2 GHz. As a result, a closed current loop is formed due to the reverse current. Similar to a magnetic dipole, the magnetic field effect between the opposite current is enhanced. The incident electromagnetic wave energy is weakened by the lossy graphene film according to Joule's law. Besides, it can be seen that the surface current on the top three layers at 8.2 GHz is much larger than that at 2.6 GHz, which forms a stronger magnetic resonance and energy damage. Corresponding to the reflection coefficient, the magnitude of S11 is -14.8 dB at 2.6 GHz while reaching -21.5 dB at 8.2 GHz as shown in Fig. 2(a). The surface current direction of the top two lossy layers at 16.4 GHz is opposite to that of the bottom lossy layer. A magnetic resonance occurred between the top three lossy layers, as shown in Fig. 6(c). Similarly, two closed current loops are formed in the top three lossy layers at 19.4 GHz, as shown in Fig. 6(d). As the resonant frequency increases, the closed current loops move to the upper layer gradually. Therefore, the absorption bandwidth could be further broadened by increasing the number of layers.
Fig. 6. Surface current distribution of radar absorber at different resonant frequencies. (a) 2.6 GHz. (b) 8.2 GHz. (c) 16.4 GHz. (d) 19.4 GHz.
Furthermore, an equivalent circuit model (ECM) is put forward to analyze the ultra-wideband absorption response. As shown in Fig. 7(a), the top and bottom lossy layer could be equivalent to a series RLC respectively, and the middle lossy layer is equivalent to a cascade of two series RLC due to the two square rings. Besides, the IRSL could be equivalent to a capacitance due to the low-pass performance. The PVC and foam are equivalent to transmission lines with the same length of thickness and a characteristic impedance of Zn, $\left( {{Z_n} = {\eta_0}/\sqrt {{\varepsilon_{rn}}} } \right),\; $n = 1, 2, where ${\eta _0}$ is free space wave impedance. According to transmission line theory and the equivalent circuit, the impedance of each layer could be expressed as follows:
Fig. 7. Equivalent circuit analysis. (a) ECM of the proposed structure. The values of components: R1 = 507.9 Ω, R21 = 421.6 Ω, R22 = 733.6 Ω, R3 = 148.9 Ω; L1 = 3.79 nH, L21 = 14.97 nH, L22 = 3.58 nH, L3 = 1.56 nH; C0 = 0.02 pF, C1 = 0.16 pF, C21 = 0.22 pF, C21 = 0.12 pF, C3 = 0.12 pF. (b) Comparison of input impedance and (c) Reflection coefficients from full-wave simulation and ECM.
Figure 7(b) shows the real and imaginary parts of Zin in ECM and commercial electromagnetic solver (HFSS), The results show that $Re ({Z_{in}})$ closes to free space wave impedance 377 Ω and $Im({Z_{in}})$ is far less than that and close to zero in the absorption band. Therefore, the input impedance of the proposed absorber is well matched to the free space wave impedance around 2∼22 GHz and a good absorption is achieved. The S11 of the equivalent circuit are basically consistent with the simulation results by HFSS simulation software, as shown in Fig. 7(c).
3. Experimental verification
3.1 Preparation and printing of conductive ink
To obtain the low sheet resistance and printable graphene conductive ink, we used liquid property-stable graphene electric aqueous slurry (GEAS) as conductive filler and purified it by drying at 100° for 1 h. In addition, carbon black (CB) was used to avoid the agglomeration of graphene because it can distribute on the surface of graphene or between graphene layers. Conversely, the graphene sheets can bridge CB particles and promote the formation of conductive paths in printed films. It is noticed that only a small amount of CB was used as conductive filler because the conductivity of CB is far less than graphene and excessive CB is disadvantageous to reduce the sheet resistance. Due to the hydrophilicity of graphene, the conductive ink was printed on a PET substrate with a contact angle of 76°, a thickness of 0.075 mm, and a dielectric constant of 3.5 which have a negligible effect on the performance of the absorber because it is thin enough comparing with the operating wavelength.
The fabrication process of graphene/CB ink was divided into three steps. Firstly, weigh the required components in Table 1 and pre-mix them in a magnetic mixer. After pre-mixing, the mixture needs to be re-mixed and defoamed by a planetary mixer/deaerator (MAZERUSTAR KK300SSE) at a speed of 2000rpm for 4 minutes. Then, disperse the graphene/CB ink ultrasonically by the ultrasonic cleaner (031ST) for 30 minutes to ensure the CB and graphene flakes are evenly dispersed without aggregation. Finally, the prepared graphene/CB ink for screen-printing is evenly printed on PET and dried in an oven at 100° for 2 h.
Table 1. Materials used in graphene ink
Due to the inevitable sheet resistance unevenness in the manual screen-printing process, a sheet resistance sensitivity analysis has been done. As shown in Fig. 8(a), when the sheet resistance of each layer is the same (RS) and varies from 30 Ω/sq to 40 Ω/sq, the absorption performance remains stable. The 90% absorption bandwidth does not change much (3.5%) which is tolerable. Since the pattern line width of the lossy graphene layer is far smaller than the probe distance of a 4-point probe station, the sheet resistance uniformity of a square pattern with a dimension of 5 × 5 mm is measured in Fig. 8(b), The sheet resistance at five locations, upper left, upper right, lower left, lower right and middle is 30.82 /29.94/31.92/ 31.26/29.56 Ω/sq, respectively. Besides, Fig. 8(c) shows the scanning electron microscopy (SEM) image of the printed graphene film. It can be seen that a small amount of carbon black is evenly distributed around the lamellar graphene which forms a conductive path.
Fig. 8. (a) Simulation reflection coefficient of the three lossy layers with the same sheet resistance RS. (b) Sheet resistance measurement. (c) SEM of the printed graphene film.
3.2 Measurement results and discussion
To verify the simulation results, a plane prototype of 25 × 25 array with an overall dimension of 300 × 300 mm and a conformal prototype of 20 × 25 array with a radius of 100 mm are fabricated and measured as shown in Figs. 9(a) and (b). The reflection coefficient is measured using the free-space method inside a microwave anechoic chamber. A pair of broadband horn antennas connected with the vector network analyzer (Anritsu MS46322A) are used to transmit and receive the EM energy. The distance between the antenna and the absorber satisfied the far field condition as shown in Fig. 9(c). The time gating domain in the vector network analyzer is set to minimize the influence of noise waves. A metal plate of equal size is used to normalize the reflection coefficient. Subject to the working frequency of the horn antenna, the measurement is carried out from 2 GHz to 18 GHz. The measured S11 of the conformal prototype is shown in Fig. 9(d), it can be seen that the conformal prototype could achieve an effective absorption and the measured S11 are less than -10 dB under TE and TM modes. It is noted that due to the bending surface, the incidence angles at different positions are different and the reflection coefficient curve is no longer smooth. Besides, the electric field direction of TE mode is parallel to the generatrix of the bending surface and the TM mode is perpendicular to that, which results in the measurement differences between TE and TM modes. For the plane prototype, the comparisons between measured and simulated results of TE/TM polarization under different incident angles are shown in Figs. 9(e)-(g). The results show that the measured reflection coefficients agree well with simulation results.
Fig. 9. (a) Plane prototype. (b) Conformal prototype. (c) Measurement environment. (d) The measured S11 of the conformal prototype. (e) Comparison of S11between full-wave simulation and measurement for plane prototype under normal incidence. (d) TE polarization oblique incidence. (e) TM polarization oblique incidence.
To demonstrate the infrared stealth characteristics of the proposed IRSL, a thermal camera (Fotric 225 s) working in the 8–14 µm range is used to measure the thermal radiation characteristics. As shown in Fig. 10(a), pieces of PET, ITO, and paper with the same area are placed on the heating table as reference samples. When the temperature of the heating table rises to 68 °C, the infrared imaging of each sample is shown in Fig. 10(b). It can be seen that the temperature of the paper is the highest, PET is next, and ITO film is the lowest which is almost undetectable. The temperature of the proposed IRSL is close to that of ITO which represents a low emissivity and it is not easily detected by infrared devices. According to (5), the emissivity of the IRSL ${\varepsilon _2}$ could be calculated by the measured temperature [42]:
where ${T_r}$ represents the temperature of IRSL measured by the thermal camera (41.8 °C), ${T_0}$ is the heating table temperature and ${T_a}$ is the ambient temperature (27 °C). Therefore, the emissivity of the proposed IRSL is 0.34 which is close to the calculated result of 0.27 according to (1). Furthermore, two pieces of IRSL with the same area are placed on the heating table and heated to the same temperature. Then the two samples are left on the heating table at the same height and one of them is curved artificially. The infrared imaging of them are shown in Fig. 10(c). It can be seen that the temperature of plane IRSL is 26.3 °C and that of the bending one is 26.4 °C. Therefore, the bending shape has little effect on the infrared radiation of the samples which could be ignored.Fig. 10. (a) Photographs of different samples. (b) Thermal imaging photos of different materials. (c) Thermal imaging photos of plane and bending IRSL. (d) Fourier transform infrared spectrometer. (e) Measured IR emissivity of the IRSL.
Besides, according to Kirchhoff's law, the emissivity is equal to the absorptivity of infrared light with the same wavelength at the same temperature. Therefore, the infrared emissivity could be calculated by (6):
where A represents the absorptivity, R represents the reflectivity and T is the transmissivity of the IRSL. The reflectivity and transmissivity from 8 µm to 14 µm are measured by a Fourier transform infrared spectrometer (Thermo Fisher IS 50) as shown in Fig. 10(d), and the calculated emissivity is shown in Fig. 10(e). It shows that a low IR emission of about 0.35 is achieved from 8 µm to 14 µm corresponding to the calculation.Table 2 lists the concerning performances of the proposed structure and other radar-infrared bi-stealth metamaterials. It can be seen that the proposed structure expresses the widest absorption bandwidth (165.4%) while keeping a low IR emission (0.35). At the same time, due to the flexible graphene lossy layers and dielectric layers, it is easy to conform to the detected object surface which expresses broad application prospects in multispectral stealth applications.
Table 2. Performance comparison of the previous radar-infrared bi-stealth structures
4. Conclusion
In summary, an ultra-wideband multilayer flexible radar-infrared bi-stealth structure is proposed. All the layers are designed based on flexible materials and the proposed structure is easy to conformal. In the radar band, ultra-wideband absorption is achieved by introducing magnetic resonance between the lossy layers. The 90% absorption band is from 1.96 GHz to 20.72 GHz with a fractional bandwidth of 165.4%. A miniaturized unit is achieved with a period of 0.079 λl, and the oblique angle incidence response is up to 45° for TE mode and 60° for TM mode, respectively. A low-pass infrared shielding layer is loaded on top of the radar absorber to realize IR stealth. The emissivity of the IRSL is as low as 0.35 from 8 µm to 14 µm which expresses broad prospects in IR stealth. A plane and a bending prototype are fabricated and measured, respectively. The screen-printing technology is adopted to print the graphene resistive films and the measurement results agree well with the simulation.
Funding
National Natural Science Foundation of China (62071357, 62171348, U19A2055); National Key Research and Development Program of China (2021YFA1003400); 111 Project; Fundamental Research Funds for the Central Universities.
Disclosures
The authors declare no conflicts of interest.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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