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Abstract
A photonic-assisted scheme for spread spectrum communication signals generation is proposed and demonstrated in this article. The spreading sequence and the baseband data codes are modulated on the photonic link by electro-optic modulators, and the spread spectrum process is completed through stream processing on the analog microwave photonic link. By combining optical frequency comb and injection locking technologies, the carrier frequency of the communication signals can be tuned over an ultra-broadband range of 3-39 GHz. In the proof-of-concept experiments, spread spectrum signals at 3 GHz and 6 GHz are obtained with a spread factor of 31. The analysis results indicate that the generated signals possess excellent reconfiguration, anti-interference, and anti-interception properties. Overall, our proposed scheme offers a flexible photonic architecture with significant potential in the application of ultra-broadband covert communication systems.
© 2023 Optica Publishing Group under the terms of the Optica Open Access Publishing Agreement
1. Introduction
Spread spectrum communication [1,2], as an attractive communication technology, has been widely used in various fields of military and civilian communication due to its excellent performances such as anti-interference, anti-multipath fading, high spectrum utilization, low intercept probability, and multiple access accommodation [3,4]. It has become the most effective measure to improve the stability and security of communication systems in the electronic countermeasures environment.
The process of generating spread-spectrum communication signals requires two steps: multiplying the spread-spectrum codes with the information data codes and loading them onto the carrier. In order to achieve a large spreading gain, a broad spreading bandwidth is needed, accordingly, a large carrier frequency is necessary. In traditional spread-spectrum systems, the spread-spectrum process is typically handled by the digital units or radio frequency (RF) front-ends, which results in broadband requirements for both digital and analog devices. Firstly, in the digital system, broadband signal processing may dramatically increase the processing time and computational resources, which may pose challenges in terms of efficiency [5]. Secondly, as for analog electrical systems, the wideband components inevitably increase the cost of the system and may distort the transmitting signal due to the inconsistent response and dispersion effect [6]. As a result, the limited bandwidth capacity of electrical devices ultimately restricts the processing gain of the system, thereby impacting the level of communication covertness that can be achieved.
Compared to electrical approaches for generating spread spectrum waveforms, photonic techniques offer wider bandwidth, better immunity to electromagnetic interference, and compatibility with radio-over-fiber technology [7–12]. In previous studies [13–18], photonic generation schemes of phase-coding microwave signals have been demonstrated with extremely simple and stable structures based on different types of modulation. In these schemes, the optical continuous wave serves as a carrier to load RF signals. Taking advantage of the large carrying bandwidth of lightwave, wideband phase coding can be generated. However, the processing of RF signals, such as the spectrum spreading, still relies on digital calculations. The advantages of high-throughput and real-time computing of photonic processing have not been involved. Additionally, in order to obtain a flexible frequency tuning range, a widely tunable RF synthesizer or optical filter is indispensable in the above schemes, which suffer from drawbacks such as large size, weight, power consumption (SWaP), and limited tunability. In conclusion, to the best of our knowledge, the photonic scheme for reconfigurable broadband spread spectrum communication signals generation has not been reported.
In this paper, we propose a scheme for spread spectrum communication signal generation assisted by optical frequency comb (OFC). The OFC [19–26] demultiplexing technology based on optical injection locking [27,28] provides a power-equalized coherent optical local oscillator with ultra-broadband tuning capability ingeniously. Additionally, spread spectrum codes and data codes are loaded on two optical carriers with different frequencies, respectively. Employing multiplication through analog microwave photonic links, the spread spectrum process is able to be completed in real time. Compared to the conventional digital spread-spectrum communication systems or novel photonic communication systems, the architecture proposed by this work exhibits two unique advantages:(1) The spectrum spreading process is implemented through microwave photonics analog links without consuming computational resources, thus, possessing the characteristics of low-latency and broadband. (2) Combining OFC and injection locking technologies, the carrier frequency of the generated signal features an ultra-wide frequency tuning range and excellent power flatness.
2. Principle
2.1 Architecture of concept
Figure 1 illustrates the schematic diagram of the proposed architecture. A lightwave from a tunable main laser (TML) is firstly split into two paths by a 3 dB optical coupler (OC). The lightwave in the upper branch is modulated by spread-spectrum codes through the Mach-Zender modulator (MZM). The spread-spectrum codes are generated from an arbitrary waveform generator (AWG). The optical carrier in the lower branch is sent to an OFC generator which is composed of cascaded phase modulators (PM), an intensity modulator (IM), and low-frequency microwave driving sources. The repetition frequency of the OFC is equal to the frequency of the microwave sources and theoretically tunable. Afterward, the OFC is sent to two cascaded distributed feedback semiconductor (DFB) lasers via circulators. Two variable optical attenuators (VOA) are employed to set the appropriate input optical power of the DFB lasers. The injection process forces the DFB laser to phase lock one of the comb lines whose frequency is closest to its resonance frequency and suppresses the other comb lines. The locking operation, therefore, has the effect of selecting and amplifying the desired comb line. Compared to the single DFB laser injection locking, cascade injection is able to further reduce the magnitude of the unwanted comb lines and provide better spurious suppression. Subsequently, communication information such as video, image, and voice are converted into bipolar data codes with a specific rule and loaded on the inject-locked comb line in the same way as the spread-spectrum codes. Then, the spread-spectrum codes in the upper path and data codes containing information in the lower path are combined in a 2${\times} $2 symmetric OC in order to realize multiplication. Finally, spread-spectrum signals are generated after photoelectric conversion through a balanced photodetector (BPD). It is worth noting that the DFB lasers are operating at a fixed frequency in our scheme, and the comb line switching is achieved by tuning the center frequency of the OFC, the same as tuning the frequency of the TML.
Fig. 1. Schematic diagram of the proposed photonic scheme to generate reconfigurable spread-spectrum communication signals. TML: tunable main laser; MZM: Mach-Zender modulator; IM: Intensity modulator; PM: Phase modulator; VOA: variable optical attenuator; CIR: circulator; DFB laser: distributed feedback semiconductor laser; BPD: balanced photodetector; OC: optical coupler; AWG: arbitrary waveform generator.
2.2 Evolutions of spectra and RF waveforms
The process for generating the signal is illustrated by the evolutions of spectra and temporal waveforms in Fig. 2. The labels in Fig. 2(a) represent their same location in Fig. 1. The output lightwave of TML is a single-frequency signal [see Fig. 2(a)①], which can be given by: $\textrm{E}$TML$(t )\propto \textrm{exp}({\textrm{j}2\mathrm{\pi }{f_0}\textrm{t}} )$. After passing through a 3 dB OC, one of the branches is modulated by spread-spectrum codes through MZM. The bias voltage is set for minimum transmission point operation. The spread-spectrum codes here are bipolar m-sequences with a bit duration of Ts, having a chip rate of Bs = 1/Ts [see Fig. 2(b)A]. Bs also denotes the main beam bandwidth of the spread-spectrum codes in the frequency domain [see Fig. 2(a)④]. After modulation, the output signal of the upper MZM in Fig. 1 is:
${V_\pi }$ is the half-wave voltage of the MZM. ${A_s}$ and m(t) are the peak amplitude and bipolar m-sequence codes, respectively. In the case of the small signal condition, Eq. (1) can be approximated to:
Fig. 2. Principle of the spread-spectrum communication signals generation illustrated by the evolutions of spectra and temporal waveforms. (a) Evolution of Spectra. (b) Evolution of RF waveforms.
Here, n is half of the total number of comb lines that provide for injection locking and $\Delta f$ is the repetition frequency of OFC [see Fig. 2(a)②]. As a result of the locking condition, the specified kth comb line is selected and amplified by DFB as an optical carrier [see Fig. 2(a)③], which is given by:
Similar to Eq. (2), after modulating by the bipolar data codes, the output of the MZM from the lower arm is:
After the combination, the optical field is converted to the electrical domain by a balanced photodetector (BPD). Hence, the photocurrent response can be represented by:
It can be seen from Eq. (7) that the generated spread-spectrum communication signals at a tailored carrier frequency can be tuned by adjusting k or $\Delta f$. Meanwhile, the communication data codes $\textrm{s}(\textrm{t} )$ multiply the m-sequence codes $\textrm{m}(\textrm{t} )$ through the analog microwave photonic link [see Fig. 2(b)C]. Thus, the narrowband data spectrum [see Fig. 2(a)⑤] is spread to a wider one [see Fig. 2(a)⑥], in which the spread factor/gain is defined by G = Tc/Ts. This may be understood intuitively by the fact that the faster the spread coding, the wider the bandwidth, and the larger the spread factor, the more covert the communication system. Since the spread codes m(t) here is a pseudo-random sequence, which has a statically noise-like spectrum with low power spectral density (PSD), the data codes are then encrypted by the coding sequence and undetectable by undesired eavesdroppers.
2.3 Reconfigurability and anti-interception
Under this scheme architecture, the format and the rate of the baseband data codes, the method of spread spectrum coding, and the setting of spread gain can be flexibly changed in the digital domain and transmitted by AWG, and the carrier frequency can be changed by the selection of different comb line (kth) of the OFC. In conclusion, we provide a computation-free, real-time, and reconfigurable spread spectrum communication signal generation scheme assisted by OFC.
The decoding of the spread-spectrum communication signals can be accomplished by digital signal processing or analog photonic processing. Due to the binary value (±1) property of the m-sequence, m(t)·m(t) = 1. Only the users who possess the accurate decoding sequence key, m(t), could simply perform decoding by multiplying the received signal with the coding sequence m(t) after proper down-conversion and synchronization. It should be noted that before transmission, the generated spread-spectrum communication signals are usually further buried under the intentionally inserted or natural background noise in order to improve invisibility. Then, after multiplying the received signal with the coding sequence key m(t), communication baseband data s(t) is recovered, at the same time, noise is expanded into a spectrum-spread low-PSD noise-like signal, which may be filtered out through a narrow bandpass filter. Therefore, the interference can be suppressed.
3. Experiment and results
3.1 Experimental setup
A proof-of-concept experiment was carried out in Fig. 3. A TML (NKT photonics) provided a lightwave at ∼1551 nm with a tuning accuracy of 2.4 pm. Next, the lightwave was adjusted to 16 dBm and divided into two branches via a 50:50 OC. Then, the upper branch lightwave was propagated to an MZM (KG-AM-15-10 G) with 10 GHz 3 dB bandwidth. The bias of the MZM was controlled by an automatic control module, which was set at the MIN point. The bipolar m-sequences with different bit durations were generated from an AWG (KEYSIGHT M8195A) with 32 GSa/s sampling rate and loaded on the MZM above. As for the lower branch, the TML served as the center frequency to generate OFC. Three PMs and one IM driven at the frequency of 3 GHz, were used to generate dozens of comb lines with the frequency interval of 3 GHz. The IM was biased at the quadrature point for flattening the spectrum. The number of comb lines can be increased by cascading more PMs. Afterward, the OFC was injected into two cascaded slave DFB lasers with the output power of ∼10 dBm and fixed at 1551.18 nm, corresponding to ∼193.40 THz. The VOA (THORLABS, VOA50PM-FC) in front of each DFB was tunable in order to achieve the suitable power for a large residual comb line suppression ratio. Then, the MZM in the lower arm has identical specifications as the upper one, which was used to load the low-speed information data codes from the same AWG. Finally, signals from two branches were combined in a 2 × 2 symmetric OC and then sent into a BPD (Finisar, BPDV21 × 0R) with a 3-dB bandwidth of 33 GHz and a responsivity of 0.45A/W. An optical spectrum analyzer (Apex, AP2040) with a resolution of 0.04 pm and an oscilloscope (Keysight, 10GSa/s, 6 GHz Bandwidth) were employed to observe the optical spectra and the waveforms of generated signals, respectively. In addition, the electrical spectra were measured by a spectrum analyzer (R&SFPH) with a maximum measurement frequency of 26.5 GHz.
Fig. 3. Experimentalsetup of the proposed spread spectrum communication signals generation system. TML: tunable main laser; optical frequency comb generator; MZM: Mach-Zender modulator; IM: Intensity modulator; PM: Phase modulator; VOA, variable optical attenuator; CIR, circulator; DFB laser: distributed feedback semiconductor laser; BPD: balanced photodetector; OC: optical coupler; AWG: arbitrary waveform generator.
3.2 OFC generation and locking
The optical spectra of OFC and different comb lines injection locking results are plotted in Fig. 4. As shown in Fig. 4(a), subject to the maximum tolerable voltage (26 dBm) of the PM, the generated electro-optic OFC provides about 27 lines with 3 GHz frequency spacing within a flatness of around 20 dB. Injection locking consists of four main steps: Firstly, we adjusted the drive current and the temperature control systems of the two DFBs to stably keep their wavelength at a precise value. Then, the wavelength of the TML was coarsely adjusted so that the wavelength of the specified comb line approached that of the DFB. Thirdly, the wavelength of the TML was finely tuned to ensure that the detuning frequency of the locking comb line and DFB was within the limits of 200 MHz. Finally, we slightly tuned the VOA until the output spectrum gradually entered a stable locking state. The whole process of injection locking can be observed by the optical spectrum analyzer. When the injection is not locked, the output spectra lines of DFB are relatively chaotic, with large noise and wide line width. When the injection is locked, the output spectral lines are narrow and clear, the noise disturbance disappears, and the state is relatively stable. As the DFB lasers are operating at a fixed frequency in our scheme, the locking comb line switching is achieved by tuning the center frequency of the OFC, the same as tuning the TML. When the wavelength of TML is set to 1551.18 nm, the DFB lasers are locked to the center comb line. The locking comb line switching can be achieved by changing the wavelength of the TML with a ∼ 0.024 nm (3 GHz) step. Figure 4(b) compares the optical spectra between cascade and single locking. It can be seen that a single DFB injection locking still leaves many useless comb lines, although the residual comb suppression ratio achieves 31 dB. Through cascade locking, the suppression is improved to 44 dB, ensuring negligible crosstalk between channels. More importantly, the output power of the desired comb line reaches ∼10 dBm, which is amplified by 30 dB through injection locking without any external optical amplifier. Figure 4(c) presents the spectra in which the DFBs are locked to the 1st, 4th, 8th, and 13th comb lines with the average residual comb suppression ratio of 40 dB, and the red arrows mark the location of the center comb line. Although the optical power of the last few channels is much lower than that near the center comb line, the injection locking can still be successfully carried out. According to Eq. (7), the carrier frequency of the final generated communication signals can achieve the range from 3 GHz to 39 GHz with the tuning step of 3 GHz. It means that the proposed scheme allows for up-conversion across several RF bands.
Fig. 4. Spectra of OFC and different comb line injection locking. (a) The spectrum of electro-optic OFC. (b) Spectra of single DFB locking and cascade DFB locking. (c) Spectra of different comb lines injection locking. (The red arrow marks the center line of the original OFC before injecting it into the DFB).
3.3 Reconfigurability of the generated signals
Firstly, the reconfigurability of the generated signal is demonstrated. Figure 5 illustrates the generation process of a spread spectrum signal centered at 6 GHz. The cascaded DFB lasers were locked to the second comb line serving as a carrier. The m-sequence was generated with the bandwidth of Bs =1452 MHz and the period number of 15 in one cycle [see Fig. 5 (a)]. It means that the original baseband data spectrum will be extended by a spread factor G of 15. A binary baseband data code with a 96.8 MHz bit rate was generated by the AWG and modulated on the carrier after amplifying by an RF amplifier. Figure 5 (b) shows the partial waveform of the spread spectrum communication signal, the phase mutation point of the sinusoidal signal with the frequency of 6 GHz is visible clearly. The spectra of the baseband signal and generated spread spectrum signal were recorded with a resolution bandwidth (RBW) of 100 kHz in Fig. 5 (c) and Fig. 5 (d). After spreading, the bandwidth of the main lobe changed from 96.8 MHz to 1452 MHz, leading to a wider spectrum. It also implies that the signal can be transmitted with a lower PSD.
Fig. 5. The generation process of a spread spectrum signal centered at 6 GHz with a spread factor G of 15. (a) The waveform of m-sequence and data codes. (b) Waveform of the spread spectrum signal. (c) Spectrum of data codes before spreading. (d) Spectrum of the spread spectrum signal
It should be noted that, in our architecture, the modulation rate of the spreading codes can be increased to ∼ 40 GHz, only limited by the bandwidth of the commercial modulators, thus the spreading factor can reach several hundred when the rate of the data codes is fixed at the order of 100 MHz. Accordingly, if a broadband OFC [29,30] is adopted, a higher carrier frequency is able to be realized, even extending to the terahertz frequency band. Limited by the measurement bandwidth of the oscilloscope, the waveform of the generated signal with carrier frequency higher than 6 GHz and larger spread factor is not shown. Nevertheless, their spectra were recorded by a spectrum analyzer with a measurement range up to 26.5 GHz. Figure 6 presents the spectra of spread spectrum signals with different carrier frequencies and spread factors. The data codes keep bit rates of 64.5 Mbps, 63.4 Mbps, and 96.8 Mbps successively, and the carrier frequencies are 12 GHz, 18 GHz, and 24 GHz. Accordingly, the spread factors are 31, 63, and 15, respectively. The above experimental results demonstrated the reconfigurability of our scheme. To demonstrate flat power output at different carrier frequencies, the electrical spectra of generated spread spectrum signals centered from 3 GHz to 24 GHz with a spread factor G of 31 are shown in Fig. 7. Although impacted by factors such as the responsiveness of photoelectric detection and the transmission loss of the cables, the output flatness can reach ∼3 dB. This confirms that the proposed spread spectrum signal generator has multi-band tuning capability with equalization output.
Fig. 6. Theelectrical spectra of original baseband data codes before spreading and spread spectrum signals and with different carrier frequencies and spread factors. (a) (b) 12 GHz carrier frequencies with a spread factor G of 31. (c) (d) 18 GHz carrier frequency with a spread factor G of 63. (e) (f) 24 GHz carrier frequency with a spread factor G of 15.
Fig. 7. The measured electrical spectra of generated spread spectrum signals centered at 3 GHz to 24 GHz with the spread factor G of 31.
3.4 Performance for anti-interception capability
Nest, the performance for anti-interception capability is verified. The generated spread spectrum signals containing 100 data codes with carrier frequencies of 3 GHz and 6 GHz were sampled by an oscilloscope and decoded through digital processing. The decoding process consists of synchronization and de-spreading. The generated signals were preferably synchronized with the decoding key, m(t), and then multiplied with it. After the bandpass filtering process, the narrowband original waveforms of the data codes were recovered and the constellation diagrams are given in Fig. 8. Figure 8(a) shows that the signals with the right decoding key (for corporation communication) are well in agreement with the original transmitted data codes and the constellation in Fig. 8(b) perfectly shows the features of the BPSK signals. While, the constellation of generated signals without decoding (for corporation communication distributes around the center point, exhibiting a weak random-noise-like feature [see Fig. 8(c)]. It proves that the signals generated by our system perform well in terms of anti-interception performance.
Fig. 8. The decoding results of the generated signal with a 6 GHz carrier and spread factor G of 15. (a) Recovered codes waveform compared with transmitted data codes waveform. (b) Constellation diagram of right decoding. (c) Constellation diagram without right decoding.
3.5 Performance for interference suppression
Finally, the performance for interference suppression of the generated signal is tested. Primarily, we compared the decoding results under the same baseband data rate and different spread factors in the case of only the inherent transmission noise of the link existing. The baseband data rate was fixed at 64.5 Mbps, and the spread factors were set at 15 and 31. Correspondingly, the bandwidth of the spread spectrum signals were 968 MHz and 2 GHz. The carrier frequencies of the transmitted signal were adjusted to 3 GHz and 6 GHz respectively. The constellation diagrams for decoding the transmitted signals are shown in Fig. 9. No matter whether the carrier frequency is 3 GHz or 6 GHz, the constellation of decoding signals becomes denser as the spread factor increases, showing a better interference suppression performance. This is because the link noise factor is deterministic, and the larger spread factor leads to lower PSD of noise lever, therefore less distortion of useful signals after decoding and filtering processing. It should be noted that when the transmission bandwidth is fixed, there is a trade-off between baseband data rate and spread gain. Thus, adequate spread gain should be chosen to balance the large capacity and low interference suppression. In order to further test the anti-interference ability of the generated signals, random noise with different power and occupying the same bandwidth as the transmitted signals were added through digital post-processing to simulate the effect of interfering. The performance of the generated signals is characterized by bit error rate (BER) measurement of the decoding codes at different signal-to-noise ratio (SNR) levels. As illustrated in Fig. 10(a), the BER decreases as the SNR increases, and at the same noise level, a higher spread gain corresponds to the lower BER. When the spread gain is 15 and the SNR is -6 dB, the BER can be reduced below 10−3. At the same SNR level, when the spread gain increases to 31, zero-BER decoding can be achieved. Meanwhile, the waveforms of signal and noise are shown in Fig. 10(b). The signal can be completely buried in the noise background, thus, the anti-interference covert communication is realized.
Fig. 9. The constellation diagrams for decoding the transmitted signals when the baseband data rate is fixed at 64.5 Mbps. (a) The spread factor G is 15 and the carrier frequency is 3 GHz; (b) The spread factor G is 31 and the carrier frequency is 3 GHz. (c) The spread factor G is 15 and the carrier frequency is 6 GHz; (d) The spread factor G is 31 and the carrier frequency is 6 GHz.
Fig. 10. The performance for interference suppression of the generated signals. (a) BERs as a function of SNRs with spread spectrum gain of 15 and 31. (b) The temporal waveforms of the generated signal signals and the strong white noises under SNR of −6 dB.
4. Conclusion
In summary, a photonic-assisted generator for spread spectrum communication signals aided by OFC and injection locking technologies has been proposed and demonstrated. Thanks to the gain equalization of OFC demultiplexing, the carrier frequency of the generated signal features an ultra-broadband frequency tuning range from 3 GHz to 39 GHz, avoiding the use of a widely tunable RF synthesizer. The process of spreading spectrum is realized through stream processing on the microwave photonic link, which does not consume computing resources and time. Although limited by the measuring equipment, the signal waveforms with the carrier frequency of 3 GHz and 6 GHz and the spreading factor up to 31 are recorded experimentally, proving the feasibility of the scheme. The constellation diagrams of decoding codes reveal the favorable anti-interception performance of generated signals. Moreover, further analysis indicates that the data codes can be efficiently hidden into the noise, and correctly recovered with a satisfactory BER performance. Furthermore, the system also has the advantages of flexible reconfigurability, arbitrary multi-band operation, and viability for photonic integration [31–35]. We believe it has great potential in the application of ultra-broadband and multi-band covert communication systems, especially in the field of low probability of detection communication.
Funding
National Natural Science Foundation of China (62075240, 62201595); National Key Research and Development Program of China (2020YFB2205804); Post-Graduate Scientific Research Innovation Project of Hunan Province (CX20210073).
Disclosures
The authors declare no conflicts of interest.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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