Channelized photonic-assisted deramp receiver with an extended detection distance along the range direction for LFM-CW radars

Jiyao Yang; Jiyao Yang; Ruoming Li; Ruoming Li; Zhenwei Mo; Zhenwei Mo; Jingwen Dong; Wangzhe Li; Wangzhe Li; Wangzhe Li

doi:10.1364/OE.386819

1. Introduction

Due to the increased requirement on range resolution, which is inversely proportional to the bandwidth of transmitted radio frequency (RF) signal, radars with capabilities of generation and processing RF signals with large bandwidths are required [1,2]. Although technology keeps improving, the capability to handle RF signals with large bandwidths is still limited by electronic devices such as analog-to-digital converters (ADCs), RF mixers, and digital signal processors (DSPs). When the transmitted signal is a linear frequency modulated (LFM) waveform, which is canonical in radars, the deramp reception or stretch processing provides a bypass by converting echoes into intermediate frequency (IF) signals whose frequencies are proportional to the distances between the targets and antennas [1,3]. Conventionally, the deramp reception is performed by RF mixers, which are made by bipolar junction transistor, field effect transistors and Schottky diodes. When using the octave-spanning to evaluate an RF mixer, RF mixers are not able to operate over one octave-spanning bandwidth, which pose a limitation of f_max<2f_min on instantaneous bandwidth of input signal, where the f_max and f_min are the maximum and the minimum frequencies of the input signal. Accompany with the development of electronic devices, the microwave photonics (MWP) technologies are also investigated in radar systems [4–11] to enhance the instantaneous bandwidth. The wideband deramp reception based on MWP technologies [12–14], especially, photonic-assisted RF mixing [14], has been reported in several works [5–10]. However, in deramp reception, the processing bandwidth, which is limited by the achievable sampling rates of the ADCs, constrains the distance along range direction of the scene to be imaged, which is called as swath depth [1].

In the paper, a novel photonic-assisted deramp receiver with an extended swath depth is proposed. By introducing the channelization technique [15–21] into the photonic deramp receiver, echoes from the targets located in a particular distance region along the range direction are deramped and digitized in a fixed detection unit. Then, a scene with a large swath depth can be detected using a group of parallel detection unites. The proposed scheme consists of an optical signal generation unit, an optical local oscillator (OLO) generation unit, and several parallel detection units. In the signal generation unit, one OFC with a free spectrum range (FSR) of F_sig is modulated on a dual-polarization quadrature phase shift keying (DP-QPSK) modulator with one polarization being intensity-modulated by the echoes and the other being intensity-modulated by the RF-reference signal. In the OLO generation unit, the other OFC with an FSR of F_OLO, is modulated by the RF-reference signal on a null-biased Mach-Zehnder modulator (MZM). Then the optical signal and the OLO are power split into a group of detection units. In each detection unit, a dual-polarization optical bandpass filter (DP-OBPF) is applied to select optical echo and the optical RF-reference from the orthogonally polarized optical signal. While in the OLO path, an OBPF is used to pick up the optical RF-reference. Then the filtered optical signal is detected with the filtered OLO in a polarization and phase diversity coherent receiver (PPDCR). Note that, beating results of each PPDCR, which include the mixing product of the optical echoes and the OLO and the mixing product of optical RF-reference and the OLO, contain common phase items, and the common phase items of the deramp signal in a detection unit can be cancelled with digital signal processing [10,22]. Since the FSRs of two OFCs are slightly different, the beating results of detection units shift in frequency with a step corresponds to the difference of the FSRs of two OFCs. By digitizing the beating results of all detection units with low-speed high-resolution ADCs, wideband echoes can be channelized during the deramp processing. As the processing bandwidth of the deramp reception is proportional to the swath depth, the swath depth can be extended by using more detection units. The performances of the proposed photonic deramp receiver are evaluated through a series of experiments.

2. Principle

The schematic diagram of the proposed system is shown in Fig. 1(a). The OFC generator, which is shown in Fig. 1(b), is built by an MZM followed by a phase modulator (PM) and a polarization-maintaining erbium-doped fiber amplifier (PM-EDFA). The generated OFC with 2N-1 comb lines can be expressed as

(1)$${E_{OFC}}(t) = E\sum\limits_{n ={-} N}^N {\exp ({j2\pi {f_o}t + j2n\pi Ft} )}$$

where f_o is the carrier frequency of the incident light wave, E is the amplitude of electronic field of each comb line, and F is the FSR of the OFC. In the optical signal generation unit, the generated carrier-OFC with an FSR of F_sig is fed into the DP-QPSK modulator shown in Fig. 1(c). In the DP-QPSK modulator, the carrier-OFC is split into two branches and sent to two QPSK modulators, on which the split carrier-OFCs are single-sideband suppressed-carrier (SSB-SC) modulated by the echoes and the RF-reference signal respectively. Assuming the transmitted LFM-CW signal of one period can be written as:

(2)$${S_{TX}}(t) = V\cos ({2\pi {f_c} t + k\pi {t^2}} )$$

where V, f_c, and k are the amplitude, center frequency, and the chirp-rate of the transmitted signal respectively. For a point target, the echo can be expressed as:

(3)$${S_{RX}}(t) = {V_{echo}}\cos [{2\pi {f_c}({t - \tau } )+ k\pi {{({t - \tau } )}^2}} ]$$

where τ = 2r/c is the round trip time of the echo, in which r is the distance between the target and the antennas. The optical signal at the output of the DP-QPSK modulator can be expressed as:

(4)$$\begin{aligned} \left[ \begin{array}{l} {{E_{Pol - X}}}\\ {{E_{Pol - Y}}} \end{array} \right] &={E_1}\sum\limits_{n ={-} N}^N {\exp \left( {j2\pi {f_o}t + j2n\pi {F_{sig}}t + j\frac{\pi }{2}} \right)} \times \\ &\left[ {\begin{array}{@{}l@{}} {j\cos \left\{ {{\beta_{echo}}\cos [{2\pi {f_c}({t - \tau } )+ k\pi {{({t - \tau } )}^2}} ]+ \frac{\pi }{2}} \right\} + \cos \left\{ {{\beta_{echo}}\cos \left[ {2\pi {f_c}({t - \tau } )+ k\pi {{({t - \tau } )}^2} + \frac{\pi }{2}} \right] + \frac{\pi }{2}} \right\}}\\ {j\cos \left[ {{\beta_{ref}}\cos ({2\pi {f_c} t + k\pi {t^2}} )+ \frac{\pi }{2}} \right] + \cos \left[ {{\beta_{ref}}\cos \left( {2\pi {f_c}t + k\pi {t^2} + \frac{\pi }{2}} \right) + \frac{\pi }{2}} \right]} \end{array}} \right] \\ &={E_1}\exp ({j2\pi {f_o}t} )\sum\limits_{n ={-} N}^N {\exp ({j2n\pi {F_{sig}}t} )} \times \left[ {\begin{array}{@{}l@{}} {\sum\limits_{l ={-} \infty }^{ + \infty } {{J_{4l - 3}}({{\beta_{echo}}} )\exp \{{j({4l - 3} )[{2\pi {f_c}({t - \tau } )+ k\pi {{({t - \tau } )}^2}} ]} \}} }\\ {\sum\limits_{l ={-} \infty }^{ + \infty } {{J_{4l - 3}}({{\beta_{ref}}} )\exp [{j({4l - 3} )({2\pi {f_c}t + k\pi {t^2}} )} ]} } \end{array}} \right] \\ &\approx {E_1}\exp ({j2\pi {f_o}t} )\sum\limits_{n ={-} N}^N {\exp ({j2n\pi {F_{sig}}t} )} \times \left[ \begin{array}{@{}l@{}} {{J_1}({\beta_{echo}})\exp \{{j[{2\pi {f_c}({t - \tau } )+ k\pi {{({t - \tau } )}^2}} ]} \}}\\ {{J_1}({\beta_{ref}})\exp [{j({2\pi {f_c} t + k\pi {t^2}} )} ]} \end{array} \right] \end{aligned}$$

where E₁ is the amplitude of each comb line of the carrier-OFC of the optical signal. In Eq. (4), β_echo=πV_echo/V_π and β_ref=πV_ref/V_π are the modulation indexes, where V_echo and V_ref are the amplitude of the echo and the RF-reference signal respectively, and V_π is the half-wave voltage of the QPSK modulators. As we can see from the Eq. (4), the echo and the RF-reference signal are replicated in optical frequency domain with a repetition interval of F_sig. The spectrum of the optical signal is shown in Fig. 2(a).

Fig. 1. (a) The schematic diagram of the proposed receiver. (b) The configuration of the OFC generator. (c) The structure of the DP-QPSK modulator. (d) The structure of the PPDCR. LNA, low-noise amplifier; EC, electronic coupler; OC, optical coupler; SG, signal generator; PS, phase shifter; PR, polarization rotator; PBC, polarization beam combiner; PBS, polarization beam splitter; BPS, beam power splitter; SIG, the signal input of PPDCR; LO, the OLO input of PPDCR; DP-EDFA, dual-polarization erbium-doped fiber amplifier.

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Fig. 2. The spectrum diagram of the optical signal and the OLO. (a) The spectrum of the optical signal. (b) The spectrum of the OLO.

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In the OLO generation unit, the carrier-OFC with an FSR of F_OLO is modulated by the RF-reference signal on a null-biased MZM. The generated OLO can be expressed as:

(5)$$\begin{aligned} {E_{OLO}} =&{E_2}\sum\limits_{n ={-} N}^N {\exp \left( {j2\pi {f_o}t + j2n\pi {F_{OLO}}t + j\frac{\pi }{2}} \right)} \times \\ &\sum\limits_{l ={-} \infty }^{ + \infty } {{{({ - 1} )}^l}{J_{2l - 1}}({{\beta_{ref}}} )\exp [{j({2l - 1} )({2\pi {f_c} t + k\pi {t^2}} )} ]} \\ \approx &- {E_2}\sum\limits_{n ={-} N}^N {\exp \left( {j2\pi {f_o}t + j2n\pi {F_{OLO}}t + j\frac{\pi }{2}} \right)} \times \\ &{J_1}({{\beta_{ref}}} )\{{\exp [{j({2\pi {f_c} t + k\pi {t^2}} )} ]+ \exp [{ - j({2\pi {f_c} t + k\pi {t^2}} )} ]} \} \end{aligned}$$

where E₂ is the amplitude of each comb line of the carrier-OFC of the OLO. Again, the RF-reference signal is replicated in optical frequency domain with a repetition interval of F_OLO. The spectrum of the OLO is shown in Fig. 2(b).

In each detection unit, the optical signal is fed into a DP-OBPF to select optical sidebands of a copy of the echo and RF-reference in dual-polarizations. The signal at the output of the DP-OBPF can be expressed as:

(6)$$\begin{aligned} \left[ {\begin{array}{c} {{E_{Pol - X}}}\\ {{E_{Pol - Y}}} \end{array}} \right] &={\alpha _1}{E_1}\exp [{j2\pi ({{f_o} + m{F_{sig}}} )t} ]\times \\ & \left[ {\begin{array}{l} {{J_1}({\beta_{echo}})\exp \{{j[{2\pi {f_c}({t - \tau } )+ k\pi {{({t - \tau } )}^2}} ]} \}}\\ {{J_1}({\beta_{ref}})\exp [{j({2\pi {f_c} t + k\pi {t^2}} )} ]} \end{array}} \right] \end{aligned}$$

where α₁ is the insertion loss of the DP-OBPF, and m, which is in the range of -N to N, is the designated serial number of comb lines. Another OBPF in the detection unit is used to select an optical replica of the RF-reference from the OLO. The optical RF-reference selected by the OBPF should be closest in wavelength to the DP-OBPF selected optical signal, so that the beating result of the filtered optical signal and the OLO can fall into the IF band and be digitized by IF-ADCs. Note that, if the OLO in the proposed scheme is also generated by a QPSK modulator in SSB-SC mode, the system would be able to operate in a frequency span wider than one octave. The signal at the output of the OBPF can be expressed as:

(7)$$\begin{aligned} {E_{OLO}} = &- {\alpha _2}{E_2}\exp \left[ {j2\pi ({{f_o} + m{F_{OLO}}} )t + j\frac{\pi }{2} - j\varphi (t)} \right] \times \\ &{J_1}({\beta _{ref}})\exp [{j({2\pi {f_c} t + k\pi {t^2}} )} ] \end{aligned}$$

where α₂ is the insertion loss, φ(t) is the relative phase between the optical signal and the OLO which is caused by the difference between two optical paths and disturbance from the environment, and m, which is in the range of -N to N, is the designated serial number of comb lines. Then the filtered optical signal and the OLO are fed into the PPDCR. The outputs of the PPDCR in the mth detection unit can be expressed as:

(8)$$\begin{aligned} {X_I} &={A_X}\cos [{2\pi ({k\tau + m\Delta f} )t + 2\pi {f_c}\tau - k\pi {\tau^2} + \varphi (t)} ]\\ {X_Q} &={A_X}\sin [{2\pi ({k\tau + m\Delta f} )t + 2\pi {f_c}\tau - k\pi {\tau^2} + \varphi (t)} ]\\ {Y_I} &={A_Y}\cos [{2\pi m\Delta f t + \varphi (t)} ]\\ {Y_Q} &={A_Y}\sin [{2\pi m\Delta f t + \varphi (t)} ] \end{aligned}$$

where Δf ≡ F_OLO - F_sig is the difference between the FSRs of two carrier-OFCs. We can see from the Eq. (8), the outputs of all parallel detection units contain the same information, except that a frequency difference of Δf is introduced to adjacent detection units. As a result, echoes corresponding to a fixed scene are converted to a series of similar IF signals with different center frequencies at the output of the parallel detection units.

Assuming the bandwidths of ADCs are wide enough, the digitized signal of both I and Q paths of each detection unit can be used to form complex quantities

(9)$$\begin{aligned} {K_X} &\equiv {X_I} + j{X_Q} = {A_X}\exp [{j2\pi ({k\tau + m\Delta f} )t + j({2\pi {f_c}\tau - k\pi {\tau^2}} )+ j\varphi (t)} ]\\ {K_Y} &\equiv {Y_I} + j{Y_Q} = {A_Y}\exp [{j2\pi m\Delta f t + j\varphi (t)} ] \end{aligned}$$

after that, by performing

(10)$$\begin{aligned} {K_{XY}}(t) &= {K_X}(t)\exp \{{ - j\arg [{{K_Y}(t)} ]} \}\\ &= {A_X}\exp [{j2\pi k\tau t + j({2\pi {f_c}\tau - k\pi {\tau^2}} )} ] \end{aligned}$$

where arg(·) is the function to obtain the polar angle of a complex number. The common items including φ(t), and mΔf can be canceled, while the information of the echoes is reserved. We can see, optical RF-reference signal in one polarization of the orthogonally polarized optical signal, corresponding to the Pol-Y in Eq. (6), acts as an optical reference signal which is phase correlated to the optical echoes.

In actual, the bandwidths of ADCs are limited, and a series of parallel IF-ADCs with relatively narrow receiving windows are used to digitize the outputs of parallel detection units. Particular portion of the mixing products of the echoes and the RF-reference is digitized by an IF-ADC in fixed detection units. After the digital processing shown in Eq. (10), the results of all detection units are stitched to recover the information of echoes. Consequently, a scene with an extended swath depth, can be detected in deramp reception.

3. Experiment and results

The proposed photonic deramp receiver is experimentally evaluated. Figure 1(a) shows the experimental setup. An arbitrary waveform generator (AWG, Keysight M8190A) is used to generate an LFM-CW signal with a center frequency of 5 GHz and a bandwidth of 2 GHz. The corresponded resolution along the range direction is 7.5 cm. The pulse repetition interval (PRI) of the transmitted LFM-CW signal is 2 µs, which means the chirp rate is 1×10¹⁵ Hz/s. The generated LFM-CW signal is amplified by an LNA and split into two parts with one used as the RF-reference signal and the other used as the transmitted signal. The transmitted signal has a power of 13 dBm and is sent to the antenna. Then, the RF-reference signal is split into two parts with a power of 10 dBm in each part. Then the split RF-reference signals are fed into the optical signal generation unit and the OLO generation unit respectively. To evaluate the system, a signal generated by the AWG, which simulates echoes from a pair of point targets distributed along range direction with distances of 9 m and 16 m, is employed. For the simulated echoes, the power of the echo from the near target is set to be 10 dB larger than that of the far target for equal radar cross-section (RCS) [1]. In the receiver, a CW light generated by a 1550 nm laser (TeraXion PureSpectrum-NLL) with a power of 18 dBm is equally split into two OFC generators. In each OFC generator, a flat carrier-OFC with an FSR of F is generated by modulating the CW lights with coherent RF signals on cascaded modulators as shown in Fig. 1(b). The frequency of the RF signal applied on the MZM is 2F, and the frequency of the RF signal applied on the followed PM is F. The RF signals with frequency of F and 2F are generated by clock synchronized RF signal generators (Rohde & Schwarz ZVA40 with four test ports and ZVA-K6 software option), as shown in the Fig. 1(b). Specifically, four signal sources of the ZVA-40 are synchronized, and RF signals with frequencies of F_sig, 2F_sig, F_OLO, and 2F_OLO are generated by the four signal sources of ZVA-40. The FSRs of the generated carrier-OFCs in the optical signal generation unit and the OLO generation unit are 17.5 GHz and 17.58 GHz respectively, and the difference between the FSRs of two carrier-OFCs is 80 MHz. The spectra of the carrier-OFCs are measured by an optical spectrum analyzer (OSA, Yokogawa, AQ6370D) and shown in Fig. 3(a), the comb lines of the carrier-OFCs are designated with a series of numbers. In the signal generation unit, the generated carrier-OFC with the FSR of 17.5 GHz is modulated on a DP-QPSK modulator (Fujitsu, FTM7977) by the simulated echoes and the RF-reference signal. The simulated echoes have been amplified by an RF chain, which is built by two cascaded LNAs with a total gain of 60 dB, followed by the receiving antenna. In the OLO generation unit, the carrier-OFC with the FSR of 17.58 GHz is modulated by the RF-reference signal on a null-biased MZM (EOSPACE, AX-DS5-20). The optical spectra of the optical signal and the OLO are shown in Fig. 3(b) and Fig. 3(c) respectively. The proposed receiver is experimentally evaluated with two detection units. In the first detection unit, a DP-OBPF is used to select the + 1 order optical sidebands of the No. 0 comb line of the orthogonally polarized optical signal, and an OBPF is used to pick up the + 1 order optical sideband of the No. 0 comb line of the OLO. Then the filtered optical signal and OLO are fed into the PPDCR respectively. In the second detection unit, the + 1 order optical sidebands of No.1 comb line of orthogonally polarized optical signal and the + 1 order optical sideband of the No.1 comb line of the OLO are selected respectively. In each detection unit, the four outputs of the PPDCR are digitized by a 4-channel oscilloscope (OSC, Keysight DSO-X 92004A) with a sample rate of 1 GSa/s and filtered in the digital domain. The OSC with a digital filter is used to substitute a band-pass filter (BPF) followed by an IF-ADC in a real receiver. Since the output of the PPDCR contains lots of noise generated by switched-mode power supply, the passband of the digital filter is set to 120 MHz to 220 MHz, corresponding to a swath depth of 12 m to 27 m for the first detection unit and 0 m to 15 m for the second detection unit. The total swath depth of the demonstrated deramp receiver is 27 m.

Fig. 3. (a) The spectra of the carrier-OFCs. (b) The spectrum of the orthogonally polarized optical signal and the frequency responses of the DP-OBPF of the first and the second detection unit. (c) The spectrum of the OLO and the frequency responses of the OBPF of the first and the second detection unit.

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The spectra of the K_X of the first and the second detection units are shown in Fig. 4(a) and Fig. 4(d) respectively, and the pass-band of the first and the second detection units are highlighted in purple and yellow respectively. As shown in Fig. 4(a) and Fig. 4(d), the echoes of the two targets can be observed in two detection units with the echo from a fixed target observed in a particular detection unit. The spectra of the K_Y of the two detection units are also shown in Fig. 4(b) and Fig. 4(e) respectively. A frequency difference of 80 MHz introduced by the difference between the FSRs of the two carrier-OFCs can be observed from the spectra of K_X and the spectra of K_Y. The K_XY of two detection units are calculated with Eq. (10), and the spectra of calculated K_XY are shown in Fig. 4(c) and Fig. 4(f) respectively. From Fig. 4(c) and Fig. 4(f) we can see the common items in the phase of the K_X and K_Y, such as the 80MHz frequency difference and the phase noise φ(t), are canceled. The distances of the point targets are calculated as 10.04 m and 17.05 m. The constant difference in distance around 1.04 m from the ideal value, which are 9 m and 16 m respectively, is attributed to the lengths of RF cables in the system.

Fig. 4. The experiment results with the simulated echoes. (a) The spectrum of K_X of the first unit. (b) The spectrum of K_Y of the first unit. (c) The spectrum of K_XY of the first unit. (d) The spectrum of K_X of the second unit. (e) The spectrum of K_Y of the second unit. (f) The spectrum of K_XY of the second unit.

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After that, an experiment using two trihedral corner reflectors (TCRs) with the same RCS of 14.7 dBsm is also performed. The distances of the TCRs are set to 9 m and 16 m respectively. The transmitted LFM-CW signal and the sample rate of the 4-channel OSC are the same as that in the experiment of the simulated echoes. The spectra of the K_X of the first and the second detection units are shown in Fig. 5(a) and Fig. 5(d) respectively, and the pass-band are highlighted with background-colors. The powers of the peaks in spectrum, which are formed by the echoes of TCR₁ and TCR₂, are 12 dBm and 5 dBm respectively. The power difference between the two peaks is 7 dB, which is close to the theoretical value of 10 dB. The 3dB derivation can be attributed to the directivity of the TCRs. The spectra of K_XY of both detection units are shown in Fig. 5(c) and Fig. 5(f), from which we can see the phase noises and the frequency differences of the FSRs included in all beating results are canceled. The distances of the TCRs can be read as 11.79 m and 18.85 m respectively. The result matches well with the real condition when considering the RF cables with total length around 1.79 m.

Fig. 5. The experiment results with a pair of static TCRs. (a) The spectrum of K_X of the first unit. (b) The spectrum of K_Y of the first unit. (c) The spectrum of K_XY of the first unit. (d) The spectrum of K_X of the second unit. (e) The spectrum of K_Y of the second unit. (f) The spectrum of K_XY of the second unit.

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4. Conclusion

A channelized photonic deramp receiver for LFM-CW radar is proposed and experimentally demonstrated. In the proposed system, optical-frequency-comb-based photonic channelization technique is integrated into photonic deramp receiver to extend the swath depth of the receiver. An optical phase correlated reference signal is introduced to the channelized deramp receiver to provide a reference for information recovery. The performances of the proposed receiver are evaluated through a series of experiments.

Funding

National Key Research and Development Program of China (2018YFA0701900, 2018YFA0701901); National Natural Science Foundation of China (NFSC) (61701476, 61690191).

Disclosures

The authors declare no conflicts of interest.

References

1. M. I. Skolnik, Radar Handbook (McGraw-Hill, 2008).

2. W. L. Melvin and J. A. Scheer, Principles of Modern Radar: Advanced Techniques, (ScitechPublishing, 2012).

3. W. J. Caputi, “Stretch: A time-transformation technique,” IEEE Trans. Aerosp. Electron. Syst. AES-7(2), 269–278 (1971). [CrossRef]

4. P. Ghelfi, F. Laghezza, F. Scotti, G. Serafino, A. Capria, S. Pinna, D. Onori, C. Porzi, M. Scaffardi, A. Malacarne, V. Vercesi, E. Lazzeri, F. Berizzi, and A. Bogoni, “A fully photonics-based coherent radar system,” Nature 507(7492), 341–345 (2014). [CrossRef]

5. R. Li, W. Li, M. Ding, Z. Wen, Y. Li, L. Zhou, S. Yu, T. Xing, B. Gao, Y. Luan, Y. Zhu, P. Guo, Y. Tian, and X. Liang, “Demonstration of a microwave photonic synthetic aperture radar based on photonic-assisted signal generation and stretch processing,” Opt. Express 25(13), 14334–14340 (2017). [CrossRef]

6. Z. Meng, J. Li, C. Yin, Y. Fan, F. Yin, Y. Zhou, Y. Dai, and K. Xu, “Dual-band dechirping LFMCW radar receiver with high image rejection using microwave photonic I/Q mixer,” Opt. Express 25(18), 22055–22065 (2017). [CrossRef]

7. F. Zhang, Q. Guo, Z. Wang, P. Zhou, G. Zhang, J. Sun, and S. Pan, “Photonics-based broadband radar for high-resolution and real-time inverse synthetic aperture imaging,” Opt. Express 25(14), 16274–16281 (2017). [CrossRef]

8. A. Wang, J. Wo, X. Luo, Y. Wang, W. Cong, P. Du, J. Zhang, B. Zhao, J. Zhang, Y. Zhu, J. Lan, and L. Yu, “Ka-band microwave photonic ultra-wideband imaging radar for capturing quantitative target information,” Opt. Express 26(16), 20708–20717 (2018). [CrossRef]

9. J. Cao, R. Li, J. Yang, Z. Mo, J. Dong, X. Zhang, W. Jiang, and W. Li, “Photonic Deramp Receiver for Dual-band LFM-CW Radar,” J. Lightwave Technol. 37(10), 2403–2408 (2019). [CrossRef]

10. J. Dong, R. Li, J. Yang, and W. Li, “Photonic stretch receiver using an optical phase correlated reference signal in a coherent-detection link,” in Proceedings of IEEE International Topical Meeting on Microwave Photonics (IEEE, 2019), pp. 1–4.

11. G. Serafino, F. Scotti, L. Lembo, B. Hussain, C. Porzi, A. Malacarne, S. Maresca, D. Onori, P. Ghelfi, and A. Bogoni, “Towards a New Generation of Radar Systems Based on Microwave Photonic Technologies,” J. Lightwave Technol. 37(2), 643–650 (2019). [CrossRef]

12. R. W. Ridgway, C. L. Dohrman, and J. A. Conway, “Microwave Photonics Programs at DARPA,” J. Lightwave Technol. 32(20), 3428–3439 (2014). [CrossRef]

13. R. Li, J. Cao, and W. Li, “Comprehensive study on parallel topological photonic mixer,” in Asia Communications and Photonics Conference (IEEE, 2018), pp. 1–3.

14. Z. Tang, Y. Li, J. Yao, and S. Pan, “Photonics-Based Microwave Frequency Mixing: Methodology and Applications,” Laser Photonics Rev. 14(1), 1800350 (2020). [CrossRef]

15. S. Pan and J. Yao, “Photonics-based broadband microwave measurement,” J. Lightwave Technol. 35(16), 3498–3513 (2017). [CrossRef]

16. V. Torres-Company and A. M. Weiner, “Optical frequency comb technology for ultra-broadband radio-frequency photonics,” Laser Photonics Rev. 8(3), 368–393 (2014). [CrossRef]

17. X. Xie, Y. Dai, K. Xu, J. Niu, R. Wang, L. Yan, and J. Lin, “Broadband photonic RF channelization based on coherent optical frequency combs and I/Q demodulators,” IEEE Photonics J. 4(4), 1196–1202 (2012). [CrossRef]

18. Y. Dai, H. Yu, K. Xu, F. Yin, Y. Ji, and J. Lin, “Optical-frequency-comb-based broadband radio frequency channelization,” in Proceedings of International Conference on Optical Communications and Networks (IEEE, 2013), pp. 1–4.

19. W. Hao, Y. Dai, F. Yin, Y. Zhou, J. Li, J. Dai, W. Li, and K. Xu, “Chirped-pulse-based broadband RF channelization implemented by a mode-locked laser and dispersion,” Opt. Lett. 42(24), 5234–5237 (2017). [CrossRef]

20. Z. Tang, D. Zhu, and S. Pan, “Coherent optical RF channelizer with large instantaneous bandwidth and large in-band interference suppression,” J. Lightwave Technol. 36(19), 4219–4226 (2018). [CrossRef]

21. H. Huang, C. Zhang, H. Zhou, H. Yang, W. Yuan, and K. Qiu, “Double-efficiency photonic channelization enabling optical carrier power suppression,” Opt. Lett. 43(17), 4073–4076 (2018). [CrossRef]

22. R. Li, X. Han, X. Chen, X. Chen, and J. Yao, “A Phase-Modulated Microwave Photonic Link With an Extended Transmission Distance,” IEEE Photonics Technol. Lett. 27(24), 2563–2566 (2015). [CrossRef]

Channelized photonic-assisted deramp receiver with an extended detection distance along the range direction for LFM-CW radars

Abstract

1. Introduction

2. Principle

3. Experiment and results

4. Conclusion

Funding

Disclosures

References

Cited By

Figures (5)

Equations (10)

Optics Express