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We propose a novel fading-free direct-detection optical orthogonal frequency division multiplexing (DDO-OFDM) scheme for 100-Gb/s medium-reach transmission. In the proposed scheme, we adopts two bands spaced at 100-GHz to accommodate the same complex-valued OFDM signal. However, the signals are coupled with a pair of orthogonal optical carriers. By doing so, real and imaginary parts of the complex-valued OFDM signal can be recovered from the two bands, respectively. We also propose a cost-effective scheme to generate such DDO-OFDM signal using an optical 90-degree hybrid and an optical I/Q modulator. The advantage of the proposed method is that it is fading-free, and the electrical spectral efficiency (SE) is doubled compared to traditional direct-detection method. Finally, we experimentally demonstrated a 101-Gb/s dual-band transmission over 320-km SSMF within only 30-GHz electrical bandwidth, which is highly competitive in both capacity and cost.
© 2015 Optical Society of America
1. Introduction
With the emergence of a variety of network applications, such as HD-TV, video sharing, and cloud computing, the end-users’ demands for bandwidth increase dramatically. In the long-haul networks, coherent detection for 100-Gb/s transmission has achieved great success due to the sophisticated optical components and powerful digital signal processing (DSP) [1–3]. Now, medium and short-reach networks, such as data center interconnection with the capacity of 40-Gb/s and beyond over hundreds of kilometers are strongly desired [4, 5].
The cost is the key issue. One solution is to lower the cost of coherent detection by using low cost components and low complexity digital signal processing (DSP). C.J. Xie et al. have reported 100-Gb/s transmission over 400-km standard single-mode fiber (SSMF) using directly modulated VCSELs and coherent detection [6]. We also demonstrated a coherent detection with low complexity DSP using the PAM-4 and orthogonal frequency-division -multiplex (OFDM) formats [7, 8]. The other solution is to use the intensity modulation and direction detection (IM/DD). Such architecture has been proved cost-effective and suitable for mass application. For example, H.G. Zhang et al. have achieved 100-Gb/s transmission of PAM-4 signal using 10-Gb/s TOSA/ROSA [9]. T. Takahara et al. have demonstrated a single channel 118-Gb/s discrete multi-tone (DMT) transmission over 10-km SSMF [10]. However, the transmission distance of these works are limited under tens of kilometers, due to the severe chromatic dispersion when data rate is ≥100 Gb/s. In order to achieve higher electrical spectral efficiency (SE) and avoid the dispersion induced fading, W. Shieh et al. propose a block-wise phase switching (BPS) [11] and stokes vector (SV) direct detection [12] to transmit complex-valued OFDM signal over hundreds of kilometers. A maximum of 1-Tb/s transmission over 480-km SSMF has been demonstrated by their group [13]. Recently, we also demonstrate a 101-Gb/s transmission over 80-km SSMF using the GBS-DD scheme [14]. However, GBS-DD scheme is at the cost of a more complex transmitter structure. For the purpose of comparison, Table 1 lists the state of art direct detection methods for the short/medium reach networks.
Table 1. Comparison between state of art direct detection methods for short/medium reach networks. S-PD: single-end photodiode; B-PD: balanced photodiode
In this paper, we propose a novel fading-free direct detection method using the two orthogonal optical carriers assisted dual-band transmission, with the same electrical SE as the single-polarization coherent system. In the proposed method, two bands spaced at 100-GHz are used to accommodate the same complex-valued OFDM signal. The signals are coupled with a pair of orthogonal optical carriers. By doing so, real and imaginary part of the complex-valued OFDM signal can be recovered from the two bands, respectively. We also propose a novel architecture to generate such dual-band signal using a 90-degree optical hybrid, which is much simpler than the structure of GBS-DD method [14]. For the reception, two single-end photodiodes are employed to detect the two bands, respectively. In the proof-of-concept experiment, we have successfully achieved 101-Gb/s (net data rate) transmission of complex-valued OFDM signal over 320-km SSMF, with more than 1-dB Q-factor margin above the 7% forward error correction (FEC) threshold for each sub-band. This is an extension work of [15].
2. Principle
The schematic diagram of the proposed direct detection method using a pair of orthogonal optical carriers is depicted in Fig. 1. In the method, two bands are used to accommodate the same OFDM signal (Es). However, the signals are assisted by a pair of orthogonal optical carriers (Ec and j*Ec). For the reception, the received optical signal is de-multiplexed. Then, the two bands are detected respectively using a pair of single-end photodiodes. In such scheme, the I/Q components of complex-valued OFDM signal can be recovered from the two bands, accordingly. We also show in Fig. 1 the principal of block-wise phase switching method for comparison, in which the I/Q components are recovered from two consecutive OFDM blocks [4].
Fig. 1 Schematic diagram of the proposed direct detection method using dual-band orthogonal carriers. De-MUX: de-multiplexer.
To generate a dual-band orthogonal carriers assisted signal, we also propose a novel scheme based on the 90-degree optical hybrid, as shown in Fig. 2(a). In the scheme, two continuous waves (CWs) noted by C1 and C2, are used at the transmitter. A 90-degree optical hybrid, with transform matrix of {C1 + C2; C1 + j*C2}, is then used to mix the CWs into two parts. The first part {C1 + C2} is modulated by an optical I/Q modulator, loading the complex-valued OFDM signal. While, the second part is directly coupled with the output of the I/Q modulator. Both lengths of the upper/lower paths should be matched to guarantee the ‘relative orthogonality’ between the OFDM signal and the optical carriers. For the reception of such signal, an optical de-multiplexer is first used to separate the two bands, which are then detected by two single-end photodiode (PD). The photo currents of the pair of PD, represented as Ii and Iq, are given by
where, Es and Ec are the signal and the optical carrier, respectively. It is straightforward to reconstruct the complex-valued signal by combining Eqs. (1) and (2),As shown in Fig. 2(b), there are three parts in the resulting complex value. The first one is a DC part that can be ignored. The second part is the signal-to-signal beat noise (SSBN) that can be eliminated by the SSBN cancellation [11]. The last term presents the fading-free complex-valued signal. From Eqs. (1) and (2), we can calculate the linearly mapped real and imaginary part of Es as We define a new term I2 asThe signal power can be evaluated from Eqs. (4) and (5) asThus, we arrive atFig. 2 (a) Proposed transmitter structure based on the 90 degree optical hybrid; (b) schematic diagram of power spectrum of the received signal.
In Eq. (8), the first fourth-order component is usually very small and thus can be ignored. Finally, we obtain as
In practice, the carrier power can be estimated by using pilot subcarriers or symbols. Thus, we can obtain the estimated SSBN according to Eqs. (6) and (9). Such dual-band scheme avoids the time-division block-wise carrier phase switching [4, 11], which greatly simplifies the transmitter structure and doubles the achievable data rate. Further reduction of the cost could be achieved by employing silicon photonic integration [16].It is noted that the I/Q imbalance caused by different loss or channel response for the two bands can be eliminated by training symbols, because the channel is assumed to be static during a period of time. In addition, there is only 100-GHz spacing between the two bands. Therefore, the channel responses and losses are nearly the same for both bands, which will have little effect on the overall performance. The polarization states of the two bands at the receiver side can be different, since each band is detected individually by a single photodiode.
3. Experimental setup
Figure 3 shows the experimental setup of the proposed dual-band direct detection transmission over 320-km SSMF. Firstly, a comb generator generates multiple optical carriers spacing at 25 GHz. The comb generator is composed of a phase modulator (PM) driven by a 25-GHz RF source. By using a wavelength selective switch (WSS), two optical carriers spaced at 100-GHz (193.35THz and 193.25THz) are selected. The selected optical carriers (C1 and C2) are then mixed by a 90 degree optical hybrid. In the upper branch, {C1 + C2} is fed into another comb generator to produce 2x5 optical comb lines. Such generator is composed of an intensity modulator (IM) driven by a 12-GHz RF source. An optical I/Q modulator is used to load the complex-valued OFDM signal, which is generated by an arbitrary waveform generator (AWG). The FFT size of the OFDM signal is 256, in which 242 subcarriers are filled with 4-QAM signal. The length of cyclic prefix (CP) is 1/64 of the FFT size. The first 10 symbols are used as training symbols (TS), which is followed by 300 payload symbols. The AWG operates at 12-GS/s sampling rate. The FEC is 7% overhead, corresponding to a 8.5-dB threshold for the Q factor. The net data rate of the 5 sub-band OFDM signal can be calculated as follows: 242 (number of loaded subcarriers) / 256 × 12 (sampling rate) × 300 (payload symbols) / 310 (symbols of payloads and trainings) × 2 (bits per sample) × 5 (sub-band number) × 0.9846 (CP overhead) × 0.935 (7% FEC) = 101 Gb/s.
Fig. 3 The proof-of-concept experimental setup for the proposed dual-band direct detection transmission over 320-km SSMF. WSS: wavelength selective switch, EDFA: erbium-doped optical fiber amplifier, SW: switch, AWG: arbitrary waveform generator.
In the lower branch, {C1 + j*C2} is delayed by a tunable optical delay line to match the length with the upper path. A polarization controller (PC) is used to align the carrier’s polarization state with the signal. Then, signals from the upper and lower branch are coupled by a 3-dB optical coupler. The CSPR can be adjusted by tuning the EDFAs in both upper and lower path. Then, the signal is fed into a recirculation loop comprising of four 80-km SSMF with Raman amplification to compensate the transmission loss. At the receiver, the received signal is first amplified, as shown in the inset of Fig. 3. A WSS, which is set to 100-GHz grid, is then used to separate the two bands. Each band is direct detected by a 40-GHz bandwidth PD. After optical-to-electrical conversion, the electrical signals are sampled by a Tektronix scope of DPO73304D operating at 100 GS/s. The offline DSP includes skew calibration between the I and Q branches, SSBN cancellation with only one-iteration, chromatic dispersion compensation, window synchronization, fast Fourier transform (FFT), channel and phase estimation, constellation reconstruction and Q-factor/BER computation.
4. Results and discussion
Figure 4 shows the Q-factor versus CSPR measurements at back-to-back for the transmission of 20.2-Gb/s net data rate, using only one sub-band. The Q-factor curve without the SSBN compensation is also shown in Fig. 4 to evaluate the impact of SSBN. The optimum CSPR increases by ~2 dB if SSBN is uncompensated. This is because more carrier power is needed to increase the power ratio of the third term in Eq. (3). With SSBN compensation, the optimum Q factor is 12.8 dB. However, the optimum Q factor drops to 10.5 dB without the SSBN compensation.
Fig. 4 Q-factor versus CSPR measurements at back to back for transmission of 20.2-Gb/s net data rate using one sub-band, with/without SSBN compensation.
We further increase the net data rate to 101-Gb/s by employing five sub-bands multiplexing. The curve of Q-factor versus CSPR at back-to-back is shown in Fig. 5. In the measurement, the optimum Q factor is observed at 10-dB CSPR, and the corresponding Q-factor is 10.3 dB. Figure 6 shows the curve of Q-factor versus launch power for 101-Gb/s transmission over 320-km SSMF. The curves are measured under three different CSPR values of 6 dB, 8 dB and 10 dB. The optimum Q factor is observed at about −2dBm launch power when CSPR is 8 dB. Less than 1-dB degradation is observed compared to the back-to-back cases. Table 2 shows the optimum Q-factors of each sub-band for the transmission over 320-km SSMF. All the Q-factors are above the FEC threshold of 8.5 dB. More than 1-dB margin for each sub-band is observed. Further increase in capacity or reach can be achieved with a wider electrical frontend bandwidth or employing higher SE modulation format.
Fig. 5 Q-factor versus CSPR measurements at back to back for the transmission of 101-Gb/s net data rate with SSBN compensation.
Fig. 6 Q-factor versus launch power for the 101-Gb/s transmission over 320-km SSMF with SSBN compensation.
Table 2. Optimum Q-factor of each sub-band, measured at 101-Gb/s rate and over 320-km SSMF.
5. Conclusion
We have experimentally demonstrated 101-Gb/s transmission of complex-valued OFDM signal over 320-km SSMF using two orthogonal carriers assisted dual-band direct detection. The electrical bandwidth is within 30 GHz. Only two PDs are used for detecting the complex-valued OFDM signal. The result shows great potential for future 100G medium-reach transmission with high electrical SE and low transceiver complexity.
Acknowledgment
This work was jointly supported by the National Natural Science Foundation of China (Grant No. 61307083), and the Major Science Innovation Project of Hubei Province (Grant No.2014AAA001).
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