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Abstract
Antenna elements with a low profile and high front-to-back (FB) ratio mean that no additional reflective cavity is required when forming the array, which greatly helps to simplify and lighten the entire array system. In this paper, we enhance the FB ratio of the antenna to 35 dB while maintaining an ultra-low profile of 0.05 λ0 by attaching the broadband polarization conversion metasurfaces (PCMs) as the parasitic patches to the surface of the radiating patch. Meanwhile, the array formed by the proposed antenna is arranged in a checkerboard form, which makes it have a lower radar cross section (RCS) in the X- and Ku- bands. Even with PCMs loaded, the antenna element maintains a small size of 0.58 λ0 × 0.58 λ0, which ensures the proposed array can achieve the ± 45° beam scanning, making it suitable for the phased array. For verification, we propose a low-sidelobe array composed of the proposed antenna elements, which exhibits a low profile, high FB ratio, and broadband RCS reduction through simulation and measurement.
© 2022 Optica Publishing Group under the terms of the Optica Open Access Publishing Agreement
1. Introduction
The operation of overlaying the metasurfaces on the targets has been widely used in the field of radar cross section (RCS) reduction [1,2]. Such metasurfaces are often fabricated based on printed circuit boards, containing dielectric substrates and arrays of metal periodic structures. By arranging two groups of sub-arrays with the reflection phase difference of ±180° by the checkerboard style, the originally concentrated scattering main lobe can be eliminated, thereby realizing RCS reduction [3–5]. The polarization conversion metasurface (PCM) can maintain the reflection phase difference around ±180° in a wide frequency band, so the RCS reduction schemes based on the PCMs are widely adopted [6–8]. In recent years, many PCMs that support multiple polarized and wide angles of incidence have been born, which has brought great inspiration to the design of low RCS antennas [9,10]. As a radiator, the antenna also needs to consider its impedance characteristics and radiation characteristics when integrated with the metasurface, so the problem of reducing the RCS of the antenna faces more difficulties.
In actual equipment, the RCS reduction of the antenna array is particularly important because of its large area and it's being placed in a prominent position. Patch antennas are widely used in antenna arrays because of their simple structure and low profile [11–14]. However, when the PCM is loaded to reduce the RCS, a certain distance is usually required between the antenna and PCM, which inevitably increases the profile of the antenna and loses a major advantage [15–21]. Even if some can achieve zero-spacing, there is a problem of insufficient front-to-back (FB) ratio [22,23], and additional reflectors need to be added, which will also destroy the low-profile characteristics. In addition to the low profile, beam scanning also presents a challenge to the size of the antenna. To avoid grating lobes, the antenna elements integrated with PCM should be small enough to ensure that the spacing of the array is 0 ∼ λ0 [24,25]. However, for a better polarization conversion effect, the sub-array of PCMs is often in form of 4 × 4 units or more, and the size of the antennas integrated with PCMs in some literature will be larger than one wavelength [26–30].
The X-band and Ku-band radars are commonly used for fire control and target tracking [31]. Because of their shorter emission wavelength and higher resolution, show a great threat to smaller-sized targets (such as antennas), so the RCS reduction in X-band and Ku-band has the highest priority for antennas. Therefore, it is desirable to achieve the RCS reduction of X-band and Ku-band while retaining the low-profile characteristics of the patch antenna, and the antenna can achieve beam scanning within a certain angle range. Being able to have an excellent FB ratio (above 20 dB) without adding a reflector, which makes it show great advantages in the simplification and lightweight of the array.
This paper proposed a low-profile and low-RCS antenna and antenna array based on PCM. Compared with existing designs, it has a lower profile, higher FB ratio, and can achieve ± 45° beam scanning. The antenna element consists of a square patch and a set of 3 × 3 PCM parasitic patch arrays, with an overall size of 0.58 λ0 × 0.58 λ0 × 0.05 λ0, which satisfies the low profile and ensures achieving ±45° beam scanning. After being integrated with the PCM, the antenna obtains better impedance matching, and the FB ratio is up to 30 dB. The proposed PCM obtains a polarization conversion rate (PCR) higher than 85% in the entire X-band and Ku-band, which greatly reduces the RCS of the antenna array in the corresponding bands. In section 3, a 4 × 4 antenna array sample is designed and fabricated, simulated, and measured to verify the realization of the above functions.
2. Low-RCS antenna array with PCMS
2.1 Eye-shape PCM and antenna element
The proposed two PCM structures are mirror-symmetrical and consist of two metallic layers and a 3 mm thick F4B substrate with the dielectric constant of ε = 2.65 + 0.002i. The unit cell of the PCM consists of a slotted ring surrounding a circular patch to form an eye shape, which is shown in Fig. 1. The details of the PCMs are: g = 0.2 mm, w = 0.4 mm, T = 12 mm, H = 3 mm, r1 = 2 mm, r2 = 3.4 mm. The PCMs are simulated by CST Microwave Studio with periodic boundary conditions and Floquet ports to extract the reflection coefficients. The periodic boundary conditions along the x- and y -axis directions of the unit-cell are applied to replace the infinite array.
Figure 2(a) illustrated the simulated reflection coefficients, and a wide polarization conversion band from 8.1 to 18 GHz is obtained. In this band, the co-polarization reflection coefficient is lower than -8dB, and the PCR is higher than 85%, the PCR in the Ku-band is higher than 95%. Since type I and type II are symmetrical, the reflection coefficients of the two types are the same, but the reflection phase in cross-polarization has a perfect ±180° difference (as shown in Fig. 2(b)).
Fig. 2. (a) Reflection coefficient of co- and cross-polarization, (b) PCR and reflection phase difference between the type I and II , (c) Simulated PCRs for three metasurfaces with different structures.
The eye-shaped PCM consists of an outer split ring and an inner circular patch. Compared with the closed ring, the split ring has begun to show the potential of broadband polarization conversion. Figure 2(c) shows the polarization conversion properties of three metasurfaces (including ring, split ring, and eye shape). In general, PCMs require resonance to occur in the v/u direction shown in Fig. 2(c), ensuring that the E-fields along the v-axis and u-axis have distinct reflection phases. From the current distributions illustrated in Fig. 3 (the incident wave is vertically polarized), the resonance occurred in the vertical direction for the closed ring, and the split ring resonance along the v-axis. For the split ring, the fundamental resonance is excited at 8 GHz, and the currents on the upper and lower metals are reversed, which is the magnetic dipole resonance; 16 GHz is the high-order resonance frequency, and the currents on the upper and lower metals are in the same direction, which is the electric dipole resonance. The circular patch in the middle enhances the resonance of the Ku-band, resulting in a continuous and high PCR in the target band.
Figure 4(a) shows a probe-fed square patch antenna with detailed dimensions as follows: l = 16 mm, p = 2.7 mm, q = 0.2 mm, and h1 = 1 mm. Its impedance is shown in Fig. 4(c), and the resonance frequency is 5.65 GHz. By loading a set of parasitic patches of a 3 × 3 PCM unit on the surface of the patch, the patch antenna can achieve better impedance matching and a higher FB ratio. The distance h2 between the PCM parasitic patch and the main patch is 2 mm, and the dielectric material between all metal surfaces is F4B. Figure 4(e) and (f) show the patterns of the above two patch antennas: after the parasitic patch is loaded, the back radiation of the patch is eliminated, and the FB ratio is increased from 20 dB to 35 dB. In addition, it is verified by simulation that the impedance and radiation characteristics of the antenna are consistent regardless of whether the loaded parasitic patch is PCM type I or type II.
Fig. 4. Geometries of the proposed patch antennas. (a) Regular patch antenna, (b) Patch antenna with PCM. (c) Impedance and (d) S11 of the proposed patch antenna. Simulated radiation patterns of proposed patch antennas: (e) E-plane, (f) H-plane.
2.2 RCS reduction principle of the antenna array
The reflection-type polarization converter can change the polarization of the reflected electromagnetic (EM) wave to the orthogonal polarization of the incident wave. Figure 5(a) and (b) demonstrate the polarization conversion mechanism diagram of the proposed PCM. The two PCMs are obtained by rotating the symmetric eye-shaped patch by ±45°, and the electric field needs to be decomposed along its two axes of symmetry (i.e., the v-axis and the u-axis). Take the Type I in Fig. 5(c) as an example, the E-field of incident EM wave polarized in x-axis can be decomposed by Ei = Eiuei(-kz-ωt) + Eivei(-kz-ωt), and the E-field of reflected EM wave can be expressed as Er = Eruei(-kz-ωt+φuu) + Ervei(-kz-ωt+φvv). Among them, φuu refers to the reflection phase of the u-axis E-field component, and φvv refers to the case of the v-axis. The simulated results of PCM along the u-axis and v-axis are shown in Fig. 5(e), the reflectivity of the two components ruu and rvv are consistent and close to 1. There is a 180° phase difference between φuu and φvv in the range of 8∼18 GHz, which causes the incident and reflected waves to be in-phase on the v-axis and out-of-phase on the u-axis. The above results in the phenomenon that the E-field angle between the reflected wave and the incident wave is 90° in Fig. 5(a) and (b).
Fig. 5. Decomposition of incident and reflected E-fields: (a) type II , (b) type I. (c) Topological diagram of the proposed checkerboard PCM, (d) proposed antenna array, (e) Simulations of reflections and phases along u- and v-axes.
However, PCM changes the polarization rather than reduces the magnitude of E-field, and the total RCS will remain unchanged. In addition, when the incident wave is polarized along the u-axis or the v-axis, the polarization conversion characteristics of PCM no longer exist. The above problems can be solved by arranging two symmetrical PCMs in the form of a checkerboard. Figure 5(c) shows the proposed checkerboard surface composed PCMs of type I and type II, where each checker is composed of 3 × 3 units cells. According to the electric field reflection diagrams provided in Fig. 5(a) and (b), no matter which direction the incident wave is polarized in, the E-fields reflected by the two adjacent checkers are always reversed and cancel each other, and the total RCS would be significantly reduced.
The simulated RCS of the proposed checkerboard PCM is shown in Fig. 6(a). As a comparison, a PEC of the same size is used as the reference surface to demonstrate the reduction of RCS. The monostatic RCS has a significant reduction in the frequency band (8∼18GHz) where polarization conversion is realized. However, not all the polarization of the reflected waves is changed, and the reduction in the RCS of the checkerboard PCM is partly determined by the polarization conversion ratio (PCR). As stated in the literature [32], the RCS reduction of checkerboard PCM can be approximated by PCR as: RCS Reduction (dB) = 10log10(1-PCR). As shown in Fig. 2(b), PCM exhibits a low PCR (0.8∼0.9) in the frequency range of 9∼10 GHz, so in the RCS reduction graph of Fig. 6(b), the RCS reduction in this part is relatively small. In other frequencies, the proposed checkerboard PCM can achieve the -10 dB RCS reduction.
Fig. 6. (a) Comparison of the simulated monostatic RCS, (b) RCS reduction of the PCM and antenna array.
For the proposed patch antenna, the PCMs appear on the antenna patch as the parasitic patches, so the effect of the antenna patch on its RCS reduction capability needs to be further investigated for the antenna array. From the RCS reduction results shown in Fig. 6(b), it can be concluded that the presence of the antenna patch does not have a dramatic impact on the RCS. At 11 GHz, the peak reduction frequency of PCM, the antenna still has the best RCS reduction capability. The difference appears in 8∼10 GHz, the RCS reduction of the antenna array is weakened compared with PCM, and in 15.5∼17 GHz, the RCS reduction is enhanced. By analyzing the surface current distribution at these frequencies, we can get a glimpse of the mystery. Taking the 3 × 3 PCM and the proposed antenna element as a comparison, their surface current distributions at 9 GHz and 16 GHz are shown in Fig. 7. For the PCM at 9 GHz, the currents are mainly distributed on the outer ring of the eye shape, and the currents on the upper metal sheet are opposite to those of the bottom metal plate, so the magnetic dipole resonances are excited at this frequency. For the antenna, the currents on the parasitic patch are consistent with the PCM, while the current direction at the center of the bottom metal plate is reversed due to the antenna patch. Although the currents on the square patch and the upper metal also maintain the magnetic dipole resonances, the reverse currents are bound to weaken the resonance and polarization conversion. For the PCM at 16 GHz, the currents distributed on the outer ring are in the same direction as the currents on the bottom metal, forming the electric dipole resonances, and the currents on the inner circle are reversed from the currents on the bottom metal, forming the magnetic dipole resonances. From the current distribution of each layer of the antenna shown in Fig. 7(d), it is not difficult to conclude that the resonance forms of the outer ring and the inner circle remain unchanged, and the resonances are enhanced due to the intervention of the square patch. Therefore, the RCS reduction of the antenna array in 15.5∼17 GHz is stronger than that of the PCM, which is based on this principle.
Fig. 7. Surface current distributions for x-polarized wave incidence at (a) 9 GHz for PCM, (b) 9 GHz for proposed antenna, (c) 16 GHz for PCM, and (d) 16 GHz for the proposed antenna.
2.3 Pattern consistency and beam scanning
As mentioned above, to realize an antenna array with low RCS, antenna elements loaded with different parasitic patches should be arranged together (as shown in Fig. 5(d)). The problem arises whether the pattern of the array composed of different antennas is consistent with the array composed of the single type antennas. The problem is clarified by introducing two linear arrays (as shown in Fig. 8(a)): the regular array consists of 8 single-type antenna elements, and the mirrored array consists of two types of antenna elements arranged alternately. Figure 8(b) is the H-plane pattern of the above two arrays with the feeding of equal amplitude and phase. It can be concluded that the patterns of the two are highly consistent, and the sidelobe levels (SLL) of the arrays are lower than -13 dB under the current antenna spacing of 36 mm. The main reasons for maintaining the consistency of the two patterns are based on two points. One is that the two parasitic patches are equal in size and symmetrical, and the resonant frequencies are the same; the other is that the polarization conversion occurs outside the operating frequency band of the antenna. There is no phase difference between the EM waves radiated by two adjacent symmetric elements, so the multi-beam phenomenon similar to that in [33] will not be generated.
Fig. 8. (a) Two types of eight-element linear arrays, (b) patterns of two arrays with equal amplitude and equal-phase feeding, (c) Beam scanning of the mirrored array.
Generally, the phased array system requires four arrays to achieve omnidirectional beam scanning in the whole space, and each array should achieve a beam scan angle of ± 45°. Figure 8(c) and Table 1 illustrated the patterns and some key parameters of the mirrored array when it is fed with constant amplitude and Δφ phase shift. As the phase shift increases from 0 to 120°, the inclination of the main beam becomes larger, and the SLL is always lower than -12 dB. In the scanning angle range of ±33°, the variation of antenna’s gain and 3 dB beam width are less than 1 dB and 2°. Considering that the antenna element spacing is 36 mm (0.58 λ0), when the maximum gain direction is 45°, a grating lobe of -5 dB appears. It’s acceptable as this grating lobe level is low and outside the ± 45° angular region. Therefore, the proposed patch antenna loaded with PCM parasitic patches is suitable for phased arrays. By cooperating with the C-band phase shifter and feeding network, the antenna and feed part of the phased array can be formed.
Table 1. Beam scanning characteristics of the proposed linear array
3. Low sidelobe array and RCS reduction
3.1 Impedance and radiation properties
Based on the idea of the sidelobe suppression in [34,35], a low-sidelobe and low-RCS antenna array (as shown in Fig. 9) is proposed in this section. It has a very simplified feed network (direct connection without bending) and is coplanar with the radiating patch, which ensures the low-profile characteristic of the array. The feed network consists of four horizontal sub-array feeders and a vertical main feeder, and the feeding point is at the center of the main feeder. The antenna elements are spaced 36 mm apart, exactly close to the guided wavelength (λg) of the center frequency, which makes all elements be fed in phase. Since there are impedance transformation sections (about λg/3 length) in the feed network, the feed amplitude of each element would be different. The equivalent circuit diagram of each sub-array feeder is shown in Fig. 9(c), and the ratio of the currents in adjacent elements can be expressed as [34]: I2/I1 = (Z2/Z1)2. By adjusting the impedance transformation ratio, the amplitudes of the antenna elements obey the Chebyshev distribution, thereby realizing low sidelobe.
Fig. 9. Topological diagram of the proposed low sidelobe array: (a) PCM layer, (b) patch and feeding. (c) Equivalent circuit of the array feeder. (d) 3-D sketch of the proposed array, (e) photographs of the fabricated samples. (f) Simulated and measured S11 of the proposed antenna array.
The geometric parameters of the feed network shown in Fig. 9(b) are set as l1 = 138 mm, l2 = 14 mm, l3 = 12.5 mm, w1 = 2.5 mm, w2 = 4.5 mm, and w3 = 6 mm. The simulation and measurement results of S11 and patterns of the array are shown in Fig. 9(f) and Fig. 10, respectively, where an Agilent 8362BE network analyzer is used for S11 measurement, and a far-field micro-wave anechoic chamber is used for pattern measurement. The measurement is in good agreement with the simulation results, and the resonant frequency of the array is 5.35 GHz. Compared with the SLL of -13 dB when feeding in the same amplitude and in-phase, the SLL of the proposed low sidelobe array is reduced to -19.3 dB in the yoz-plane and -15.7 dB in the xoz- plane (refer to the coordinate system of Fig. 9(d)). The high FB ratios of 33.1 dB and 33.2 dB are achieved on these two planes synchronously.
Fig. 10. Simulated and measured patterns of the proposed antenna array: (a) yoz-plane, (b) xoz-plane.
3.2 RCS-reduction performance
The simulated bistatic scattered fields under normal& oblique incident EM waves with x-polarization and 45°-polarization are illustrated in Fig. 11. Obviously, the scattering main lobe of the array loaded with PCM parasitic patches is eliminated compared to the reference initial array. Due to the anisotropy of the feed network, the scattering properties of its two polarizations need to be investigated separately. The monostatic RCS measurements of the reference initial array and PCM array are performed in an anechoic chamber (shown in Fig. 12).
Fig. 11. Bistatic scattered fields under (a) normal incidence with x-pol. at 11 GHz, (b) normal incidence with 45°-pol. at 11 GHz, (c) normal incidence with x-pol. at 17 GHz, and (d) 45° oblique incidence with x-pol. at 11 GHz.
Fig. 12. RCS measurement setup for cylinder (a) schematic diagram of the top view (b) measured environment.
The simulated and measured monostatic RCS for two proposed arrays for x-polarization and y-polarization under normal incidence are presented in Fig. 13. The -4 dB RCS reduction achieved in 8∼10 GHz and -8 dB RCS reduction achieved in 10∼19 GHz, which is consistent with the RCS reduction capability of the antenna array in Fig. 6. Accordingly, it can be determined that the feed network of the array has a minor effect on its RCS. The average RCS reduction in the target X and Ku bands can reach -11.8 dB, which achieves the stealth performance.
Fig. 13. Simulated and measured monostatic RCS for two proposed arrays: (a) x-polarization, (b) y-polarization.
The stealth technology requires that the array have consistent RCS reduction characteristics for generally polarized incident waves. For the plane wave shown in Fig. 14(a), the angle between the electric field and the x-axis is θ, and the propagation direction follows the z-axis. Figure 14 illustrates the RCS reduction by the PCM array with the polarization angles of θ = 15°, 30°, and 45°, the RCS reduction results are consistent for the incident waves of these three polarizations, indicating that the stealth capability of the array is insensitive to the polarization angle.
Fig. 14. RCS reduction for different polarized incidence waves: (a) Simulated results, (b) Measured results.
Except for the case of vertically incident waves, for oblique incident waves, the array still has a certain stealth capability. For a flat metal target, the maximum direction of the scattered energy is the mirror direction of the incident wave (as shown in Fig. 15(a)), and the Bistatic RCS in this direction needs to be investigated. When the incidence angle φ = 15°, 30°, and 45°, the corresponding Bistatic RCS reduction results are shown in Fig. 15. For the small angle (15° or 30°), the RCS reduction results are consistent with the conclusion of vertical incidence. When the angle increases to 45°, the peak of the array's RCS reduction will diminish, but the average will still be below -5 dBsm, still with significant stealth capabilities.
Fig. 15. RCS reduction for different oblique incidence waves: (a) Simulated results for x-pol, (b) Measured results for x-pol, (c) Simulated results for y-pol, (d) Measured results for y-pol.
A comparison between the previously reported low-RCS antenna arrays and this work is illustrated in Table 2. The advantage of our work is that while ensuring the RCS-reduction in the critical frequency bands, the antenna element can still be compact, low profile, and high FB ratio, making it suitable for the various types of planar arrays such as phased arrays. In terms of stealth performance, its advantages are that it is not sensitive to the polarization angle of the incident wave, and has consistent and excellent RCS reduction characteristics for arbitrarily polarized incident waves. The RCS reduction for oblique incidence waves is also significant, and the RCS reduction ability will be weakened when the angle of incidence is greater than 45.
Table 2. Comparison of the performances among previously low-RCS arrays and this work
4. Conclusion
By loading two symmetrical PCM patches on the square patches and arranging them in a checkerboard form, a beam-scannable array with low RCS in X and Ku bands is obtained. In terms of radiation characteristics, the proposed antenna array can achieve ±45° beam scanning and a high FB ratio of 35 dB while maintaining an ultra-low profile of 0.05 λ0. In terms of scattering characteristics, it can achieve -4 dB RCS reduction at 8∼10 GHz, -8 dB RCS reduction at 10∼19 GHz, and an average reduction of -11.8 dB in X and Ku bands. In the field of antenna stealth, this work is one of the rare solutions that can simultaneously achieve the RCS reduction, beam scanning, low profile, and lightweight.
Funding
National Natural Science Foundation of China (41704176); Fundamental Research Funds for the Central Universities (2042020gf0003).
Disclosures
The authors declare no conflicts of interest.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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