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Abstract
A broadband, electrically controlled, reconfigurable, circularly polarized reflective metasurface is presented. The chirality of the metasurface structure is changed by switching active elements, which benefits from the tunable current distributions generated by the elaborately designed structure under x-polarized and y-polarized waves. Notably, the proposed metasurface unit cell maintains a good circular-polarization efficiency in a broadband range of 6.82-9.96 GHz (fractional bandwidth of 37%) with a phase difference of π between the two states. As a demonstration, a reconfigurable circularly polarized metasurface containing 8 × 8 elements was simulated and measured. The results verify that the proposed metasurface can flexibly control circularly polarized waves in a broadband, realizing beam splitting, mirror reflection, and other beam manipulations from 7.4 GHz to 9.9 GHz (fractional bandwidth of 28.9%) by simply adjusting the loaded active elements. The proposed reconfigurable metasurface may offer a promising approach to electromagnetic wave manipulation or communication systems.
© 2023 Optica Publishing Group under the terms of the Optica Open Access Publishing Agreement
1. Introduction
Compared to conventional materials in nature, metasurfaces are not limited to the intrinsic properties of their component materials. For instance, phase manipulation by conventional optical lenses relies on the spatial gradient thickness of the glass. As a comparison, metasurfaces can generate an abrupt phase difference at the interface between free space and the metasurface due to their ability to flexibly manipulate the equivalent permittivity and permeability. In addition, metasurfaces have merits such as low profile, easy fabrication, and low cost.
Most metasurfaces [1–5] manipulating electromagnetic (EM) waves in air can be divided into transmissive and reflective metasurfaces. Transmissive metasurfaces always require more layers to achieve better EM wave control to obtain extensive phase coverage. Otherwise, breaking the upper limit with a single-layer structure is difficult [6,7]. Benefiting from the metallic ground plane [8], reflective metasurfaces can use a single layer to achieve a more extensive phase range with high efficiency. Thus, various reflection-phase metasurfaces have been reported [9–11]. However, once a passive metasurface has been fabricated, it cannot be modified again, providing a static working state. As the EM environment becomes more complex, this single working state will become more limited. Therefore, dynamic phase-state controllable metasurfaces have received much attention.
Dynamic metasurfaces are generally controlled by switching the state of each active element in the array, thus obtaining different phase states and further dynamically controlling the beam [12,13]. In the past few years, researchers have reported optical control [14], temperature control [15], phase change material control [16,17], and electrical control [18,19] for dynamic phase manipulation of EM waves. However, most current active metasurfaces control linearly polarized waves [20,21], and there are fewer ways to dynamically control circularly polarized waves due to the complexity of controlling the rotation of the geometric phase elements. Some works obtain the geometric phase by loading a motor under each element to control the structure rotation [22]. Although this approach avoids quantization loss in switch-type active elements, the large size and slow rotation speed are inherent limitations. The use of liquid-metal ring-shaped resonators may pave the way to control circularly polarized waves, but the reported works have mainly focused on manipulating linearly polarized waves [23,24]. Some works obtain phase differences by orthogonally changing the direction of the current [25]. This method has a relatively fixed and complex approach to feeding and usually does not operate in a wide frequency band.
In this work, we propose an appealing method to dynamically control the circularly polarized reflection phase in a broadband based on the current distribution switching of the designed subwavelength structure. A reconfigurable circularly polarized metasurface (RCPM) is fabricated to verify the wave front manipulation capacity. The results demonstrate that two different phase states can be generated by current mode switching to control the deflection of circularly polarized waves. The proposed approach uses a smaller number of lumped elements and has the capability to control circularly polarized waves over a wide frequency band, which will be beneficial for different circularly polarized beam control applications in wireless communications.
2. Results and discussion
2.1 Concept and analytical design
The proposed RCPM is shown in Fig. 1(a). When a circularly polarized (CP) incident wave vertically impinges on the RCPM, two different reflection phases can be produced by the RCPM due to the electrically controlled state of the element. Thus, the RCPM can control the reflected circularly polarized beam as desired, for example realizing flexible beam splitting and mirror reflection. Each row of elements is connected to a voltage source by bias lines. In addition, a field-programmable gate array (FPGA) is used to dynamically control the feed voltage of each row. As a result, the RCP EM waves can be actively controlled by the RCPM with simple voltage control.
Fig. 1. Proposed RCPM and its element. (a) Schematic model of the RCPM array and its multiple electronically controlled states: beam splitting and mirror reflection. (b) Schematic model of the element.
The RCPM element is shown in Fig. 1(b). The overall structure of the RCPM element consists of two substrates, including an F4BM-2 (ε = 2.2, tan δ = 0.003) 3.2 mm top layer and an F4B (ε = 2.65, tan δ = 0.003) 0.17 mm bottom layer. The split-resonant-ring-like structure etched on the top dielectric substrate is symmetric about the diagonal, and there are two slots 90° apart on the outer ring for loading positive- intrinsic-negative (PIN) diodes. A fan-shaped stub is utilized to make the overall structure equivalent to an open circuit when feeding through the bottom layer. A metallized ground is inserted between the top and bottom dielectric substrates. In addition, a metallized hole VIA1 is used to connect the upper surface pattern with the bias line, while VIA2 is employed to connect the upper surface pattern with the ground. It is noted that all the metallized are made of copper (electric conductivity σ = 5.96e + 007 S/m). The period of the element is 11.7 mm. Other geometric parameters of the element are as follows: L1 = 1.95 mm, L2 = 0.9 mm, L3 = 0.92 mm, W1 = 0.7 mm, W2 = 2.4 mm, W3 = 0.5 mm, W4 = 0.95 mm, W5 = 0.25 mm, D1 = 1.7 mm, D2 = 3.96 mm, R1 = 2.1 mm, R2 = 2.9 mm, R3 = 3.4 mm, R4 = 4.9 mm, R5 = 3 mm, α1 = 75°, α2 = 45°, and h1 = 3.5 mm. Two MADP-000907-14020x PIN diodes (MACOM) are embedded in the top structure of the element to realize switchable phase control, and the equivalent circuit is exhibited on the top left of Fig. 1(b). The two diodes are distributed counterclockwise, that is, they cannot operate in the same state when operating normally. When the feed line is fed with positive voltage V+, PIN1 is OFF and PIN2 is ON for state 1; when the line is fed with negative voltage V-, PIN1 is ON and PIN2 is OFF for state 2.
2.2 Numerical analysis
To investigate the phase and amplitude characteristics of the proposed RCPM, we use the CST Microwave Studio software to simulate the element with periodic boundary conditions. As shown in Fig. 2(a), when an RCP plane wave is incident on the RCPM element, both state 1 and state 2 have a co-polarized reflection magnitude greater than -1 dB in the band with a fractional bandwidth of 37% between 6.82 GHz and 9.96 GHz. The cross-polarized magnitude in the band is below -15 dB, corresponding to a circularly polarized axial ratio (AR) of less than 3 dB, thus obtaining a good circular-polarization purity. The two states have a stable phase difference of approximately 180° in the frequency range, as shown in Fig. 2(b).
Fig. 2. RCPM element in the two states under incident EM waves (a) Co-polarized and cross-polarized magnitudes under an RCP incident wave. (b) Phase difference of the element in two states under an RCP incident wave. (c) Magnitude and phase difference under linearly polarized incident waves. (d) The blue curve is the cross-polarized magnitude for Model 1, the black curve is the cross-polarized magnitude for Model 2, and the red curve is the cross-polarized magnitude for the proposed model. Electric field intensity at different frequencies of the proposed model under the RCP incident wave: (e) 7.1 GHz, (f) 8.6 GHz, and (g) 9.7 GHz.
To achieve different states of circularly polarized waves, we can use vectors to deduce the states of their incident and reflected waves. Supposing that the incident wave is an RCP wave incident along the -z-axis direction, the incident field Ein can be expressed as:
The reflected fields Eref1 and Eref2 for the two states corresponding to the RCP incident wave can be expressed as:
Assuming reflectance values rx1 = ry1 = rx2 = ry2 = 1, the difference between the two reflected states of Eq. (2) can be expressed as:
Comparing Eq. (4) with Eq. (5), the following conditions are required to achieve state switching:
According to Eq. (6), if θ = π, then the phases of the orthogonal components of the two states must meet φx1 - φx2 = π ± 2nπ and φy2 – φy1 = π ± 2nπ, where n is a positive integer. That is, the phase values of the x-component and y-component of the reflected waves for the different states can be used to manipulate the reflection states as desired to obtain high reflectivity of circular polarization and a 1-bit phase difference.
In Fig. 2(c), the proposed RCPM element is separately illuminated by an orthogonal linearly polarized wave. Both state 1 and state 2 have an excellent co-polarized reflection magnitude (insertion loss is approximately -1 dB). The phase difference of approximately π between state 1 and state 2 illuminated by the same linearly polarized incident waves indicates that the element can satisfy the conditions for state switching as in Eq. (6). The helicity of the incident CP wave can be maintained after reflection [26]. In addition, we have broken down the RCPM structure to better explain the principle of the proposed element operation. The resonance peak can be adjusted to broaden the operating band by adding a perturbation structure to the basic split ring resonator Model 1. As shown in Fig. 2(d), when we etch the slot above the outer ring to change Model 1 to Model 2, we can effectively shift the resonance peak to expand the frequency band. Furthermore, by adding a smaller ring into Model 2, more resonance points can be generated over a broader frequency band. The cross-polarized magnitude curve of the proposed model, given by the red-dotted line in Fig. 2(d), shows that another high-frequency resonance point is created at 9.7 GHz. Figures 2(e)–2(g) illustrate the electric field distributions on the surface of the proposed model when PIN1 is OFF and PIN2 is ON. The electric field strength of the outer ring edge is the strongest when operating at 7.1 GHz. The electric field strength of both the outer ring and inner ring is moderate when operating at 8.6 GHz, and the electric field strength of the inner ring is the strongest when operating at 9.7 GHz. The two mutually nested rings make the element resonate over a broader frequency band, effectively generating a broad operation frequency band.
Furthermore, we analyze the current distribution of the proposed RCPM element, as shown in Fig. 3. The x-polarized and y-polarized incident waves are used to illuminate the surface of the elements. As shown in Figs. 3(a) and 3(b), when PIN1 is OFF and PIN2 is ON (state 1), at 7.1 GHz, the x-polarized wave produces the current distribution Jx1on the split ring, and the y-polarized wave produces the current distribution Jy1. In contrast, when PIN2 is OFF and PIN1 is ON (state 2), at 7.1 GHz, the y-polarized and x-polarized waves produce the current distributions Jy2 and Jx2, as shown in Figs. 3(c) and 3(d). Jx1 and Jy2 are symmetrical about the diagonal of the split-resonant-ring-like structure, and Jy1 and Jx2 are the same. In addition, Figs. 3(e)–3(l) exhibit the current distributions of the element at 8.6 GHz and 9.7 GHz, which also demonstrated that switching the PIN diodes would exchange the current distributions on the metasurface structure under x-polarized and y-polarized incident waves. Accordingly, the reflective phase difference of the structure under the orthogonal linearly polarized waves at different states still maintain a π phase difference, as observed in Fig. 2(c). That is, the different states of PIN diodes cause the switching of current distributions on the metasurface structure under the x-polarized and y-polarized incident waves, further causing the π phase difference of reflected circularly polarized waves in the two states. In addition, the overall dimensions of the element remain unchanged, which guarantees a similar reflection magnitude in the different states of the element.
Fig. 3. Current distributions of the element surface with x-polarized and y-polarized incident waves: (a, b) State 1 at 7.1 GHz. (c, d) State 2 at 7.1 GHz. (e, f) State 1 at 8.6 GHz. (g, h) State 2 at 8.6 GHz. (a, b) State 1 at 9.6 GHz. (c, d) State 2 at 9.6 GHz. State 1 represents that PIN1 is OFF and PIN2 is ON, and state 2 represents that PIN1 is ON and PIN2 is OFF.
3. Experiment results
Next, a metasurface sample consisting of 8 × 8 proposed RCPM elements (95.6 × 95.6 mm2) was fabricated and measured to verify the ability of the proposed metasurface to manipulate circularly polarized waves, as shown in Fig. 4(a). The two layers of F4BM substrates are bonded together by the solidified sheet FR28 (ε = 2.8, tan δ = 0.003, thickness of 0.1 mm). We connect the RCPM elements of each column to a voltage source through the bottom bias line and connect the ground to the middle of the positive and negative terminals of the two voltage sources. The diode has a conduction voltage of 1.35 V, which means that we need to connect the positive terminal of one voltage source to the element that requires V + and the negative terminal of the other voltage source to the element that requires V-. When the PIN diode is on, Ron = 5.2 Ω, and L = 0.05 nH. When the PIN diode is off, Roff = 15000 Ω, Coff = 0.025 pF, and L = 0.05 nH. The sample is measured in a microwave anechoic chamber. As shown in Fig. 4(b), two RCP horn antennas are connected to a vector network analyzer as the receiver and transmitter. The transmitter horn is placed 1000 mm in front of the sample and used to obtain a quasi-plane wave. The transmitter horn and the sample are fixed to the rotation equipment. When the rotation equipment is revolved, the sample is always illuminated by the incident wave from the transmitter horn at a fixed angle. The reflected wave can be received by the receiver horn at a rotation angle of ± 90°. Figure 4(c) shows the reflectivity of the RCPM sample measured by the free space method. The magnitude is large than -2.5 dB from 7 to 10.5 GHz and is approximately -1.5 dB at most operating frequencies.
Fig. 4. Schematic diagram of the sample and measurement system. (a) Fabricated RCPM sample. (b) Scattering pattern measurement system in the far-field anechoic chamber. (c) Reflectivity of the RCPM array sample and unit cell.
Compared to the proposed element, the RCPM sample reaches a similar working band but makes a blue shift. The discrepancies between the element and the array are caused by the different simulated boundaries and different incident waves.
Before testing the scattering patterns of the RCPM sample, we calculate the phase distribution of the metasurface according to the equation Φ=2π/λdsinθ. Φ is the phase shift of the metasurface, d is the distance of each element from a reference element, θ is the metasurface scan angle, and λ is the center frequency wavelength. Then, the calculated phase shift values between 0 and π are categorized as 0, corresponding to the reflected phase for element state 1, and the calculated phase shift values between π and 2π are categorized as π, corresponding to the reflected phase for element state 2.
Figure 5 depicts the measured and simulated results of the sample for different beam-forming states. The radiation patterns at 7.4 GHz, 8.6 GHz and 9.9 GHz are chosen to describe the performance of this sample in the operating frequency band. The phase distributions of the metasurface in three beam-forming states are shown in Figs. 5(a), 5(e) and 5(i). As shown in Figs. 5(b)–5(d), at the calculated scan value of ±24° at 8.6 GHz, the corresponding array state is “ππ0000ππ”, where the measured main reflection peak is oriented to approximately ±24° at 8.6 GHz, which is the same as the simulated angle. In Figs. 5(f)–5(h), the array state is changed to “0000ππππ”, and the position of the main peak is redirected to approximately ±14°, which is in good agreement with the simulated angle. For the specular state “00000000”, the transmitter is tilted by 15° for irradiation to prevent the reflected beam from being blocked by the horn. Both the simulated and measured results have a main peak position at approximately 15°, which are in good agreement, as shown in Figs. 5(j)–5(l). The deflection angle of the beam can be controlled under all three array conditions, which indicates that the deflection bandwidth is at least 7.4-9.9 GHz. Note that some discrepancies emerged in the simulated and measured results, which is caused by the nonideal measurement setup. First, the differences between the measurement and simulation environments have an impact on the consistency of the measured and simulated results. For example, the horn antenna during the measurement actually provides a quasi-plane wave, and the uneven energy distribution may affect the reflected field distribution. In addition, the energy incidentally scattered by the support bracket of the horn and the rotating equipment can be directly received by the receiver horn, thus generating additional noise. Nevertheless, a broadband circularly polarized beam reconfiguration property of the designed metasurface is achieved.
Fig. 5. Measured and simulated beam scattering results of the RCPM sample. (a), (e), and (i) Phase distributions of each column in the array represented using two colors. Normalized measured and simulated radiation patterns at 7.4 GHz, 8.6 GHz, and 9.9 GHz: (b), (c), and (d) for the state of “ππ0000ππ”, (f), (g), and (h) for the state of “0000ππππ”, (j), (k) and (l) for the state of “00000000”.
To further study the reconfigurable capacity of the proposed RCPM, we designed a 2D reconfigurable reflectarray antenna composed of 8 × 8 proposed elements. In Fig. 6(a), the 2D RCPM array is fed by a circularly polarized patch antenna at 8.6 GHz. First, the focusing length is defined as 40 mm, and the phase distribution on the surface of the metasurface is calculated by the electromagnetic simulation software. Second, the phase compensation for steering the beam to different angles is precalculated and discretized into the 1-bit phase distribution. At last, the proposed reconfigurable metasurface with the designed 1-bit phase distribution is simulated. Figure 6(b) shows the simulated 3D radiation patterns for the scanning angle of 20° in the xoz planes. The high directivity can be realized by the proposed reconfigurable reflectarray antenna. In addition, Fig. 6(c) shows the simulated radiation patterns for different scanning angles in the xoz planes. Several directive beams that scan from 0° to 50° are calculated. The largest scanning angle is observed at 47° with the predesigned angle of 50°. The 20° beam achieves the greatest gain with a value of 12.6 dBi, which increases 5.9 dB to the gain of the feed antenna. This is mainly due to the blockage of the patch antenna, where its maximum scattering should occur somewhere larger than theta = 0°. Figure 6(c) can indicate that the limit scanning range of the proposed reconfigurable metasurface for a single beam is from -47° to 47°. It is noted that the performance can be further enhanced by employing a larger array and the phase distribution optimization algorithm.
Fig. 6. Designed reflectarray antenna and simulated radiation patterns at 8.6 GHz. (a) The 3D model of the designed reflectarray antenna comprises a patch feed antenna and an RCPM array. (b) The 3D radiation pattern for a steering beam of 20°. (c) The 2D radiation pattern for steering beams from 0° to 50°.
4. Conclusions
In summary, an electrically controlled 1-bit reconfigurable circularly polarized metasurface is proposed and measured in this paper. The designed reconfigurable metasurface is demonstrated for dynamical switching of the phase state of circularly polarized reflected waves between 0 and π based on current mode switching. Dynamic circularly polarized beam control is successfully achieved over a wide frequency range from 7.4 to 9.9 GHz. In addition, the simplicity of the structure feeding method and the small number of PIN diodes make this metasurface has less processing error, more stable performance, and easier manufacturing, potentially providing a promising approach to manipulating the circularly polarized reflected waves in a broadband.
Funding
National Natural Science Foundation of China (62031006, 62205038); China Postdoctoral Science Foundation (2021M693712).
Acknowledgements
The authors would like to thank Fushun Hao in Beijing University of Posts and Telecommunications, School of Information and Communication Engineering.
Disclosures
The authors declare no conflicts of interest.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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